MIC_UNIT 4 - Copy.pptx microwave integrated circuits

18ECE304T-MICROWAVE INTEGRATED CIRCUITS
UNIT IV – Mixers and Microwave Diodes
S1- Introduction to Mixers
S2-Mixer Types, Conversion loss ,S3- SSB Mixers
S4- DSB Mixers, S5 to S7- Microwave Diodes
S8-S9 – Attenuators & Phase shifters
Dr.K. Suganthi
Assistant Professor
Department of ECE, SRM IST

S1- Introduction to Mixers
• Mixers are commonly used to multiply signals of different frequencies
in an effort to achieve frequency translation.
• The motivation for this translation stems from the fact that filtering
out a particular RF signal channel centered among many densely
populated, narrowly spaced neighboring channels would require
extremely high Q filters.
• The RF signal carrier frequency can be reduced or down converted
within the communication system.

Heterodyne Mixer
• The received RF signal is, after preamplification in a low-noise amplifier (LNA), supplied to a mixer whose task is to multiply
the input signal of center frequency f RF with a local oscillator (LO) frequency f Lo.
• The signal obtained after the mixer contains the frequencies f RF k f LO, of which, after low-pass (LP) filtering, the lower
frequency component f RF - f LO, known as the intermediate frequency (IF), is selected for further processing.
• The two key ingredients constituting a mixer are the combiner and detector. The combiner can be implemented through the
use of a 90' (or 180") directional coupler
• In addition to diodes, BJT and MESFET mixers with low noise figure and high conversion gain, have been designed up to the
X-band

Mixer- Introduction
• A mixer is capable of taking two frequencies at its input and
producing multiple frequency components at the output.
• A nonlinear device such as a diode, FET, or BJT can generate multiple
harmonics.
Figure : Basic mixer

Mixer Design
• The RF input voltage signal is combined with the LO signal and supplied to a semiconductor device with a
nonlinear transfer characteristic at its output side driving a current into the load.
• Both diode and BJT have an exponential transfer characteristic, as expressed for instance by the Shockley diode
equation.
Alternatively, for a MESFET we have approximately a square behavior:
where the subscripts denoting drain current and gate-source voltage are omitted for simplicity. The input voltage is
represented as the sum of the RF signal
This voltage is applied to the nonlinear device whose current output characteristic can be found via a Taylor series
expansion around the Q-point:

MIC_UNIT 4 - Copy.pptx microwave integrated circuits

Frequency Domain Considerations
After performing mixing the resulting spectral representation contains both
upconverted and down converted frequency components.
Typically the upconversion process is associated with the modulation in a
transmitter, whereas the downconversion is encountered in a receiver.
Lower sideband, or LSB ( w, - wLo )
Upper sideband, or USB (aw + wLo )
Double sideband, or DSB ( W, + wLo, W, - wLo ).
An interrelated issue is the problem of image frequencies mapping into the
same down converted frequency range.

Image frequency mapping
• To avoid the presence of undesired image signals that can be greater in magnitude than the RF signal, a so-
called image filter is placed before the mixer circuit to suppress this influence, provided sufficient spectral
separation is assured. More sophisticated measures involve an image rejection mixer.

Key Parameters- Mixer
• Conversion loss or gain between the RF and IF signal powers
• Noise figure
• Isolation between LO and RF signal ports
• Nonlinearity

Basic diode characteristic
I
V
+
-
I
V
Is
+
-
Id
Vd Cp
Lp
Cj
(V) Rj(
V)
V-I characteristic Equivalent circuit
 
1
)
( 
 V
s e
I
V
I  where a= q /nkT , q =charge, k=Boltzmann’s constant,
T = temperature, n = ideality factor and Is = saturation
current.
Package
components
Rs
Junction
components
Contact
resistance

Continue
Let’s say diode voltage V = Vo + u where Vo is a DC bias voltage and u is a
small AC signal voltage. We expand using Taylor series
n
n
n
R
n
a
x
a
f
a
x
a
f
a
x
a
f
a
f
x
f












)!
1
(
)
)(
(
....
)
)(
(
!
2
1
)
)(
(
)
(
)
(
1
)
1
(
2
"
'
Taylor series
By substituting, we have (x-a) = ( Vo+u -Vo)= u and the Taylor series for I(V) is
Taking f(x) = I(V), then x= Vo+u and a = Vo
 
1
)
( 
 V
s e
I
V
I   
1
)
( 

 o
V
s
o
o e
I
V
I
I 
....
2
1
)
(
2
2
2




o
o V
V
o
dV
I
d
dV
dI
I
V
I 

where and
Reminder

Continue
By substituting and Io= I(Vo) in the first derivative
 
j
d
s
o
V
s
V R
G
I
I
e
I
dV
dI o
o
1




 
 
Similarly in the second derivative, we have
  '
2
2
2
2
d
d
s
o
V
s
V
d
V
G
G
I
I
e
I
dV
dG
dV
I
d o
o
o





 

 
 
1
)
( 
 V
s e
I
V
I 
...
2
)
( '
2



 d
d
o G
G
I
V
I


Then (400)
 
1
)
( 

 o
V
s
o
o e
I
V
I
I 
where

Rectifier application
If the diode voltage consist of DC and small RF signal V = Vo + uo cos wot
where Vo is a DC bias voltage and u cos wot is a small RF signal voltage. Then
by substituting into (400)
....
cos
2
cos
)
( 2
'
2



 t
G
t
G
I
V
I o
d
o
o
d
o
o 



RF in DC out
....
2
cos
4
4
cos '
2
'
2




 t
G
G
t
G
I o
d
o
d
o
o
d
o
o 





continue
Rearrange
....
2
cos
4
cos
4
)
( '
2
'
2




 t
G
t
G
G
I
V
I o
d
o
o
d
o
d
o
o 




DC rectified current
AC harmonics current of frequency wo and 2wo. This
can be filtered off by using lowpass filter

Detector application
  t
t
m
V o
m
o 

 cos
cos
1

Modulated RF Detected RF
Modulated signal representation
where m = modulation index
wm= modulation frequency
wo= RF carrier frequency

continue
 
  ....
cos
cos
1
2
cos
cos
1
)
(
2
2
'
2






t
t
m
G
t
t
m
G
I
V
I
o
m
d
o
o
m
d
o
o






   
   
   

































t
t
t
m
m
t
m
t
t
m
t
m
m
G
t
m
t
m
t
G
I
m
o
m
m
o
m
o
m
o
m
o
o
m
m
d
o
m
o
m
o
o
d
o
o



















2
cos
2
cos
2
cos
2
2
cos
2
cos
2
cos
2
cos
2
cos
2
2
1
4
cos
2
cos
2
cos
4
4
2
2
2
'
2
2
2

Trigonometry relationship
   
 
y
x
y
x
y
x 


 cos
cos
2
1
sin
sin
   
 
y
x
y
x
y
x 


 cos
cos
2
1
cos
cos
   
 
y
x
y
x
y
x 


 sin
sin
2
1
cos
sin
   
 
y
x
y
x
y
x 


 sin
sin
2
1
sin
cos
 
x
x 2
cos
1
sin
2 2


 
x
x 2
cos
1
cos
2 2



continue
From the eq. above we have several harmonics as shown with relative
amplitude.
A
mplitude

0

m
2

m

o
-
m

o

o

m

o

m
)

o

o

m
)

o

m
)

o

m
)
1+m 2/2
2m
m2
/2
m
m
1+m 2/2
m2/4
m2/4
k
km/2
km/2
k=uoGd/(uo
2
Gd’/4)
=4/(uoa)
As linear
detector
( ~uoGd)
As squared detector
~uo
2
Gd’/4

Square-law region of diode detector
We are measuring power , thus square-law region is to be chosen since the
power measured is proportional to uo
2
. If we want to measure voltage , then
the linear region is the choice. For linear detector,we choose frequency at wo
and for square-detector at 2wo.. Using filter we can filter out the modulating
frequency wm.
logP in
(dBm)
log v out
Saturation
Noise level
Square-law region
1V
Vout = o
2
=P in
100mV
10mV
1mV
100 V
10 V
-30 -20 -10 0 10 20 30

Single-ended mixer
RF AMP MIxer
Local
Oscillator
IF AMP
Lowpass
filter
fRF
fRF
fLO
fIF
=fRF
-fLO
RF input
MIxer
Local
Oscillator
Bandpass
filter
fRF
fIF
fLO
fIF=fRF +fLO
IF input
RF AMP
Downconverter
Upconverter
The purpose of mixer is to convert either from one frequency to higher frequency
or vice versa. The advantages of conversion are (i) to reduce 1/f noise when
convert to lower frequency (ii) for easy tuning for a wide band with fixed IF and
(iii) frequency off-set between transmitter and receiver by using a single LO as in
Radar.

Simplest Single-ended mixer
•Uses nonlinearity of a diode property
•The output generated consist of frequencies spectrum dc component,
wr,wo,wr-wo, wr+wo.
•For IF, we filter out all frequencies except wr-wo.
•For upconverter, we filter out all lower frequencies and allow only wr+wo.
•Combiner either T-junction or directional coupler
•Matching network is to match the output of combiner to the diode
circuitry.
bandpass
filter
vicos( r-o)t
Matching
network
Combiner
DC bias
LO
vocos ot
vrcos rt r
, o
,
r+ o
RFC
RFC

analysis
Let’s
 2
cos
cos
2
'
t
v
t
v
G
i o
o
r
r
d 
 

Then substituting into equation (400) and we have for the second term as
 
t
v
t
t
v
v
t
v
G
o
o
or
r
o
r
r
r
d 


 2
2
2
2
cos
cos
cos
2
cos
2
'



 
 
 t
v
v
t
v
v
t
v
t
v
v
v
G
o
r
o
r
o
r
o
r
o
o
r
r
o
r
d














cos
2
cos
2
2
cos
2
cos
4
' 2
2
2
2
t
v
v r
r
RF 
cos
 t
v
v o
o
LO 
cos

an
d
DC
Figure of merit in mixer is its conversion loss, defined as
 
dB
power
output
IF
power
input
RF
available
Lc log
10


Single Balanced Mixer Circuit
Advantages
•For either better input SWR or better RF/LO isolation
•Cancellation of AM noise from LO
* Note that, although it is not shown, the diodes required biasing and
matching network.
/4
/4
Zo
Zo
Zo
Zo
2
/
Z o
2
/
Z o
Zo Zo
diode
Diode
RF
LO
LPF
LFP

analysis
Let’s
     
  t
v
v
t
v
t
v
v
t
v
t
v
o
n
o
r
r
o
o
n
o
o
r
r




cos
sin
180
cos
90
cos
)
(
1








The voltages across the two diodes of 90o
out of phase is given as
t
v
v r
r
RF 
cos
   t
t
v
v
v o
n
o
LO 
cos
)
(


an
d
Where vr<<vo and vn(t)<<vo
     
  t
v
v
t
v
t
v
v
t
v
t
v
o
n
o
r
r
o
o
n
o
o
r
r




sin
cos
90
cos
180
cos
)
(
2









Diode 1
Diode 2
Vn is a small random noise voltage

Diode current
Assuming identical diodes so that diode currents can be represented as
2
1
1 kv
i 
   
 
t
t
v
v
v
t
v
v
t
v
k
i o
r
n
o
r
o
n
o
r
r 


 cos
sin
2
cos
sin 2
2
2
2
1 




2
2
2 kv
i 

and (reverse polarity)
   
 
t
t
v
v
v
t
v
v
t
v
k
i o
r
n
o
r
o
n
o
r
r 


 sin
cos
2
sin
cos 2
2
2
2
2 





     

     
 
t
t
v
v
v
t
v
v
t
v
k
o
r
o
r
n
o
r
o
n
o
r
r
















sin
sin
2
2
cos
1
2
cos
1
2
2
2
     

     
 
t
t
v
v
v
t
v
v
t
v
k
o
r
o
r
n
o
r
o
n
o
r
r

















sin
sin
2
2
cos
1
2
cos
1
2
2
2
Dc and lower frequency bands

IF frequency band
 
    
t
v
v
v
v
v
v
k
i o
r
n
o
r
n
o
r 
 




 sin
2
2
2
2
1
 
    
t
v
v
v
v
v
v
k
i o
r
n
o
r
n
o
r 
 





 sin
2
2
2
2
2
After low pass filtering, the remaining terms are dc and IF frequency terms, thus
Written the IF frequency wi = wr- wo then the IF current is
  t
v
kv
t
v
v
kv
i
i
i i
o
r
i
n
o
r
IF 
 sin
2
sin
2
2
1 






where vn << vo . This show that the noise in the first order is cancelled by
the mixer while the desired IF signal combined in phase.

Anti parallel diode mixers
RF input
Bandpass
filter for RF
Lowpass
filter for LO
and IF
LO input
Lowpass
filter for IF
IF
output
r
o
i
 
i
r
o 

 

2
1
The LO is one-half of usual LO, I.e
The non-linearity of diode generates 2nd harmonic of LO to mix with RF(wr)
to produce desired IF. The anti parallel diode creates symmetrical V-I
characteristic that suppresses the fundamental product of RF and LO. It also
suppresses AM noise.

Double Balanced mixer
180 o
hybrid
RF input
LO input
IF
output
Zo


Single -ended mixer produces output consisted of all harmonics. The
balanced mixer using hybrid suppresses all even harmonics of the LO.
Double balanced mixer suppresses all even harmonics both LO and RF.

Image rejection mixer
3dB
power
divider
RF input
Mixer A
Mixer B
90 o
hybrid
LO
LSB
USB
IF out
90 o
hybrid
Zo
The RF with frequency wr= wo + wi will also produce the IF (wi) when
mixed with LO. The frequency produced will be USB(wr= wo + wi ) and
LSB(wr= wo - wi ) . The undesired frequency either USB or LSB is called
image frequency. The mixer can produce one single side band is used as
modulator.

Analysis
Let RF signal consist of both upper and lower sidebands
   t
v
t
v
v i
o
L
i
o
U
r 


 


 cos
cos
   t
v
t
v
v i
o
L
i
o
U
A
r 


 


 cos
2
cos
2
Then input to mixer A and B
 
   
 
o
i
o
L
o
i
o
U
B
r t
v
t
v
v 90
cos
2
90
cos
2





 



After mixing with LO, wo , The IF’s produced by mixer are.
t
kv
t
kv
v i
L
i
U
A
i 
 cos
2
2
cos
2
2


   
o
i
L
o
i
U
B
i t
kv
t
kv
v 90
cos
2
2
90
cos
2
2



 


Analysis
Both IF , then combined in the 90o
hybrid produces LSB and USB.
 
 
t
kv
t
v
t
v
t
v
t
v
k
v
i
L
i
L
o
i
U
i
L
i
U
LSB





cos
2
cos
180
cos
cos
cos
4






   

   
t
kv
t
v
t
v
t
v
t
v
k
v
i
U
o
i
L
o
i
U
o
i
L
o
i
U
USB





sin
2
90
cos
90
cos
90
cos
90
cos
4










Conversion Loss
• The FET realization allows not only for LO and RF isolation but also provides signal gain and thus minimizes
conversion loss.
• The conversion loss (CL) of a mixer is generally defined in dB as the ratio of supplied input power PRF over
the obtained IF power PIF :
• When dealing with BJTs and FETs, it is preferable to specify a conversion gain (CG) defined as the inverse of
the power ratio. Additionally, the noise figure of a mixer is generically defined as
• CG again being the conversion gain, and P, , Pnin the noise power at the output due to the RF signal input (at RF) and
the total Gise power at the output (at IF). The FET generally has a lower noise figure than a BJT, and because of a
nearly quadratic transfer characteristic, the influence of higher-order nonlinear terms is minimized.
• Instead of the FET design, a BJT finds application when high conversion gain and low voltage bias conditions are
needed

Nonlinearities
 Nonlinearities are customarily quantified in terms of conversion compression and
intermodular distortion (IMD).
 Conversion compression relates to the fact that the IF output power as a function of RF
input power begins to deviate from the linear curve at a certain point. The point where
the deviation reaches 1 dB is a typical mixer performance specification.
 Intermodulation distortion is related to the influence of a second frequency
component in the RF input signal, giving rise to distortion. T
 To quantify this influence, a two-tone test is typically employed. If f RF is the desired
signal and f is a second input frequency, then the mixing process produces a frequency
component at 2 f - f RF f f LO, where the +/- sign denotes up- or downconversion.
 The influence of this intermodulation product can be plotted in the same graph as the
conversion compression

Conversion compression and intermodulation
product of a mixer

General single-ended mixer design approach

Single-ended mixer design
 The circuit design of an RF mixer follows a similar approach as discussed when
dealing with an RF amplifier.
• The RF and LO signals are supplied to the input of an appropriately biased transistor
or diode.
• The matching techniques of the input and output side directly apply for mixers as
well. However, one has to pay special attention to the fact that there is a large
difference in frequencies between RF, LO on the input side, and IF on the output side.
• Since both sides have to be matched to the line impedance, the transistor port
impedances (or Sparameter representation) at these two different frequencies have
to be specified.
• Furthermore, to minimize interference at the output side of the device, it is
important to short circuit the input to IF, and conversely short circuit the output to RF

Single-Balanced Mixer
 The single-ended mixers are rather easy to construct circuits. The main
disadvantage of these designs is the difficulty associated with providing LO
energy while maintaining separation between LO, RF, and IF signals for
broadband applications.
 The balanced dual-diode or dual-transistor mixer in conjunction with a hybrid
coupler offers the ability to conduct such broadband operations.
 Moreover, it provides further advantages related to noise suppression and
spurious mode rejection. Spurs arise in oscillators and amplifiers due to
parasitic resonances Basic Characteristics of Mixers and nonlinearities and are
only partially suppressed by the front end.
 Thermal noise can critically raise the noise floor in the receiver.

Balanced mixer involving a hybrid coupler
Figure below shows the basic mixer design featuring a quadrature
coupler and a dual-diode detector followed by a capacitor acting as
summation point.

Balanced mixer involving a hybrid coupler
• Besides an excellent VSWR , it can be shown that this design is
capable of suppressing a considerable amount of noise .
• Because the opposite diode arrangement in conjunction with the 90"
phase shift provides a good degree of noise cancellation

Single-balanced MESFET mixer with coupler
and power combiner

Single-balanced MESFET mixer with coupler
and power combiner
• A more sophisticated design, involving two MESFETs and 90" and 180"
hybrid couplers involves a 180" phase shift.
• 180" phase shift is needed since the second MESFET cannot easily be
reversed as done in the anti-parallel diode configuration
• It is also important to point out that this circuit exhibits LO to RF as
well as LO to IF signal isolation, but no RF to PF signal isolation.
• For this reason, a low pass filter is typically incorporated into the
output matching networks of each of the transistors

Double-Balanced Mixer
• The double-balanced mixer can be constructed by using four diodes arranged
in a rectifier configuration.
• The additional diodes provided better isolation and an improved suppression of
spurious modes.
• Unlike the single-balanced approach, the double-balanced design eliminates all
even harmonics of both the LO and RF signals.
• However, the disadvantages are a considerably higher LO drive power and
increased conversion loss.
• From the figure all three signal paths are decoupled, and the input and output
transformers enable a symmetric mixing with the LO signal.

References
1. RF circuit design Theory and applications, Reinhold Ludwig, Pavel Bretchko

Gunn diode
1.Gunn diode is very much temperature dependent i.e., a
frequency shift of 0.5 to 3 MHz per °C.
2.By proper design this frequency shift can be reduced to 50 kHz
for a range of - 40°C to 70°C.
3.Other disadvantages of Gunn diode is, the power output of the
4.Gunn diode is limited by difficulty of heat dissipation from the
small chip.
5.Gunn diode is very much temperature dependent.

Applications
Gunn diode can be used as an amplifier and as an oscillator. The
applications of Gunn diode are
1. In broadband linear amplifier.
2. In radar transmitters.
3. Used in transponders for air traffic control.
4. In fast combinational and sequential logic circuit.
5. In low and medium power oscillators in microwave receivers.

AVALANCHE TRANSIT TIME DEVICES
In 1958, Read at Bell Telephone Laboratories proposed that the
delay between voltage and current in an avalanche,
together with transit time through the material, could make a
microwave diode exhibit negative resistance such devices are
called Avalanche transit time devices.
The prominent members of this family include the IMPATT and
TRAPATT diode.
1. IMPATT (Impact Ionization Avalanche Transit Time) diode as the
name suggests, utilizes impact ionization for carrier generation.
2. TRAPATT (Trapped Plasma Avalanche Triggered Transit Time)
diode is derived from the IMPATT with some modifications in the
doping profiles so as to achieve higher pulsed microwave powers
at better efficiency values.

IMPATT DIODE
1. The IMPATT diode or IMPact Avalanche Transit time diode is an
RF semiconductor device that is used for generating microwave
radio frequency signal, with the ability to operate at frequencies
between about 3 to 100 GHz or more, one of the main
disvantages is the relatively high power capability of the IMPATT
diode.
2. IMPATT Structures
There is a variety of structures that are used for the IMPATT diode
like p+nin+ or n+pip+ read evice, p+nn+, and p+in+ diode, all are
variations of a basic pn junction
IMPATT diode is semiconductor device which generate microwave
signal from 3 to 100 GHz.
In IMPATT diode, negative resistance effect phenomenon is taken
into account

• The Schottky barrier diode has a different reverse-saturation current
mechanism, which is determined by the thermionic emission of the majority
carriers across the potential barrier.
• This current is orders of magnitude larger than the diffusion-driven minority
carriers constituting the reverse saturation
current of the ideal pn-junction diode.
Schottky Diode
W. Schottky analyzed the physical phenomena involved when a metallic electrode
is contacting a semiconductor. For instance, if a p-semiconductor is in contact with
a copper or aluminum electrode.
There is a tendency for the electrons to diffuse into the metal, leaving behind an
increased concentration of holes in the semiconductor.
The consequences of this effect are modified valence and conduction band
energy levels near the interface.

Typical component values for Schottky diodes are Rs=2.. . 5 R,
C, = 0.1 . . . 0.2 pF, and Rj = 200 . . . 2 kR.
Often, the additional IRs term in is neglected for small bias currents below 0.1
mA.
However, for certain applications, the series resistance may form a feedback
loop, which means the resistance is multiplied by a gain factor of potentially
large magnitude. For this situation, the ZRs term has to be taken into account.
Equivalent circuit with values

PIN Diode
• PIN diodes find applications as high-frequency switches and variable resistors
(attenuators) in the range from 10 kohm to less than 1 ohm for RF signals up
to 50 GHz.
• They contain an additional layer of an intrinsic (I-layer) or lightly doped
semiconductor sandwiched between highly doped p+ and n+ layers.
• Depending upon application and frequency range, the thickness of the
middle layer ranges from 1 to 100 µm.
• In forward direction, the diode behaves as if it possesses a variable
resistance controlled by the applied current. However, in reverse direction
the lightly doped inner layer creates space charges whose extent reaches
the highly doped outer layers.
• This effect takes place even for small reverse voltages and remains essentially
constant up to high voltages, with the consequence that the diode behaves
similar to a dual plate capacitor.
• For instance, a Si-based PIN diode with an internal I-layer of 20 µm and a
surface area of 200 by 200 µm has a diffusion capacitance on the order of
0.2 pF.

Varactor Diode
• The PIN diode with its capacitive behavior under
reverse bias already suggests that a variable
capacitance versus voltage characteristic can be
created by a specific middle layer doping profile.
• A varactor diode exactly accomplishes this task by a
suitable choice of the intrinsic layer thickness Win
addition to selecting a particular doping distribution
ND(x) .

Pulse generation of varactor diode

Varactor diode
If we assume W = 10 micrometer and vdmm = 10 cm/s to obtain a transit time
that is equivalent to a pulse width of

Pin Diode Equivalent Circuit
Cp
Lp Rs R1
C1
PIN diode resistance
1
10
100
1000
10000
0.0001 0.01 1 100
Forward bias (mA)
RF
resistance
(Ohm)
0.1
1
10
100
1000
RF
conductance
(mOhm)
N+
I
P+
symbol

Equivalent circuits for ON and OFF states
of PIN diodes
Li
Rr
Li
Rf
Cj
Zr
Zf
ON
state
OFF
state Reverse bias will provide OFF state
Forward bias will provide ON state

Single-pole PIN diode Switches
Series
+V
R SW
C1
Diode
RFC1
RFC2
C2
Parallel
R
+V
C1
SW
RFC
Diode
C2
RF in
RF in
RF out
RF out
ON =No RF out
OFF= RF out
ON= RF out
OFF=No RF out
Note: C1 and C2 are dc block

Simplified switching circuits
Zd
Zo
Zo
Zo
+
VL
_
+
VL
_
2V o
2V o Zd
Zo
o
L
V
V
IL log
20


d
o
o
Z
Z
Z
IL



2
2
log
20
In general, the insertion loss
Series switch
Shunt switch
o
d
d
Z
Z
Z
IL



2
2
log
20
 









i
f
f
j
i
r
r
d
L
j
R
Z
C
L
j
R
Z
Z


 1
where

Example
A single-pole switch is to be constructed using a PIN diode with the
following parameters: Cj= 0.1pF, Rr= 1W, Rf= 5 W , Li= 0.4nH. If the
operating frequency is 5 GHz and Zo= 50W, which circuit (series or shunt)
should be used to obtain the greatest ratio of off-to-on attenuation?
Solution
 















6
.
12
5
.
0
7
.
305
1
1
j
L
j
R
Z
j
C
L
j
R
Z
Z
i
f
f
j
i
r
r
d



dB
Z
Z
Z
IL
d
o
o
11
.
0
2
2
log
20 


 dB
Z
Z
Z
IL
o
d
d
03
.
0
2
2
log
20 



dB
Z
Z
Z
IL
o
d
d
07
.
7
2
2
log
20 



dB
Z
Z
Z
IL
d
o
o
16
.
10
2
2
log
20 



Series switch Shunt switch
ON
OFF
state
Ratio
10.05dB Ratio
7.04dB

Other Single pole single throw PIN
Switches Configuration
50 
D1
D2
D1 D3
D2 50 
L- SPST
Switch
T- SPST
Switch
Single Pole
Single Throw
Note:
Biasing are not shown,
just diodes configuration

SPST Switches performance
Ty pe Isolation Insertion
Series 
















2
10
2
1
log
10
o
c
Z
X







o
s
Z
R
2
1
log
20 10
Shunt 






s
o
R
Z
2
1
log
20 10

















2
10
2
1
log
10
c
o
X
Z
L



































2
2
2
10
1
2
2
1
log
10
s
o
o
c
s
o
R
Z
Z
X
R
Z











 














2
2
10
2
2
1
log
10
c
s
o
o
s
X
R
Z
Z
R
T












































2
2
10
2
10
2
2
1
log
10
1
log
10
s
c
s
o
o
c
R
X
R
Z
Z
X















 



















2
10
2
10
2
1
log
10
1
log
20
c
s
o
o
s
X
R
Z
Z
R

PIN diode switching operation
Isolation Vs Diode resistance
25
30
35
40
45
50
55
60
0 1 2 3
Diode resistance (ohm)
Isolation
(dB)
By putting diodes in parallel will
reduce the total diode resistance
and thus improves isolation as
shown in graph.
AC
V
50 
Switch
50 
Source Load
Diode "OFF"
-Switch "ON"
Diode "ON"
-Switch "OFF"
Equivalent circuit
Switch Configuration
(Shunt diode)

PIN diode switch (improving isolation)
AC
V
50 
Switch
50 
Source
Load
Equivalent circuit
4
AC
2550 
1 50 
Isolation vs line length
35
40
45
50
55
60
65
70
75
80
85
0 100 200
Line length(deg)
Isolation
(dB)
Rd=1.5ohm
Rd=1ohm
Rd=0.5ohm
Isolation is maximum when the
transmission line is exactly 90o
and
the effect of diodes similar to without
transmission line when its length equal
to 0o
or 180o
.
50W

PIN diode switch(input impedance not 50W
AC
V
Rs
Switch
50 
Source
Load
4
Compensating
line
Compensating line is a 90o
transmission line to match the Rs
with 50ohm line.Rs is lower than 50 ohm.
50W

All-shunt Diode Nonreflective SPST Switch
 

Input
D3
D1 D2
D4

B1
Output


AC
V
50 
50 
Source
Load
Equivalent circuit
Diode "ON"
(Switch "ON")
Diode "OFF"
(Switch "OFF")
(Serial diode)
Isolation vs Diode capacity
0
5
10
15
20
25
30
35
40
0 0.2 0.4 0.6 0.8 1
Diode capacity, Cd (pF)
Isolation
(dB)
By putting diodes in series will
reduce the total effective series
capacity, thus increase isolation.
This is shown in graph below.

AC
V
50 
Switch
50 
Source Load
50 
4
In this case the optimum line
line is not 90o
, but depend on
the diode capacity.
Isolation vs Line length
0
10
20
30
40
50
60
70
0 50 100 150
Line length(deg)
Isolation
(dB)

Single pole double throw PIN diode switches
/4
/4
Output 1 Output2
Output 1 Ouput 2
Input
Input
Series
Shunt
Operation
•One diode is biased in low
impedance state with another
diode in the high impedance
state, so that input signal can be
switched to one output to the
other by reversing the diodes
state or biasing.
•The quarter-wave lines limit of
the shunt circuit limit the
bandwidth.

PIN diode application
(TR switch)
Transmitter
Receiver
dc supply
Antenna

D1
D2
dc block
dc block
Dc supply “ON” for transmitting. D1 and D2 will conduct,
allowing the signal from transmitter to go to antenna and
any signal go to receiver will be shorted. When dc supply
“OFF”, D1 and D2 will not conduct, thus allowing only signal
from antenna flow to the receiver.

(Reflective phase shifter)
RF input
RF output
A B
C
D




90 o Hybrid






 r
2

Switched line phase shifter
2

1

in out
 
2
1 
 

 

•Using two single pole and double throw switches to route the signal
between one of two transmission lines of difference length.
•The phase difference is . This circuit is a broadband &
reciprocal phase shifter so that it can be used as receiver as well as
transmitter.
•Disadvantages-resonance when the length is multiple of l/2 and
frequency is shifted due to diode capacitance.

(8-steps phase shifter)



D2
D1
D2
D2
D1
D1
RFC RFC RFC RFC RFC RFC
A B C

(8-steps phase shifter)
• When A, B or C is set “1” then D1 and D2 will conducted allowing the
RF go straight.
• When A,B or C is set “0” then D1 and D2 will not conducted and the
RF signal will experienced phase shift according to the length of U -
line.
• l/2 is 90o
phase shift, l/4 is 45o
phase shift and l/8 is 22.5o
phase
shift.

Switching equivalent phase shift
0 0 0 157.5o
0 0 1 135o
0 1 0 112.5o
0 1 1 90o
1 0 0 67.5o
1 0 1 45o
1 1 0 22.5o
1 1 1 0o
A B C Phase shift

PIN diode application Bridged T attenuator
D1 (R s1
)
D2 (R s2)
Zo
RF input RF output
Zo










1
1
log
20
s
o
R
Z
A 2
1
2
s
s
o R
R
Z 

where
Attenuation is small when D2 is forward biased (low impedance) and D1 is
reverse biased (high impedance). Conversely , attenuation is large when D2 is
reverse biased (high impedance) and D1 is forward biased (low impedance).
in
out
V
V
a 
Attenuation factor
is defined as

PIN diode application p attenuator
RF Input RF Output
D1 (R s1
)
D2 (R s2)
D3 (R s3)










2
3
1
log
20
s
s
R
R
A
Attenuation is small when D3 is forward biased (low impedance) and D1and D2
are reverse biased (high impedance). Conversely , attenuation is large when D3 is
reverse biased (high impedance) and D1 and D2 are forward biased (low
impedance).

(intermodulation attenuator)
+20V
OUT50/75 
R2
R4
R5 R6
IN 50/75 
R3
R1
D1
D2
Vin=0-20V
R1, R2 2.2k
R3, R4 1k
R5, R6 75ohm
D1, D2 UM9301unitrode
All capacitors are 470pF ceramic
At high input voltage and
low attenuation, D1 tends to
conduct signal.R1 and R2
set the current and isolate
the dc. D2 tends to be off.
At low input voltage and
high attenuation, D1 tends to
be off. D2 tends to bypass
the signal to ground. R3 and
R4 set the current and isolate
the dc. R5 and R6 maintain
the characteristic impedance

Input Voltage Vs Attenuation (dB)
0
10
20
0 5 10 15 20
Input Voltage (V)
Attenuation
(dB)
100MHz
200MHz
400MHz
Attenuation of signal after applying Vin for frequency 100MHz
to 400MHz

Input Voltage Vs Return loss (dB)
10
15
20
25
0 5 10 15 20
Input Voltage (V)
Return
loss
(dB)
100MHz
200MHz
400MHz
Return loss is less than 10 dB. It seem the impedance
characteristic is maintain.

Attenuator
Input
Output
Coupler
Diode
Diode
Bias
Input
Output
Coupler Coupler
Diode
Diode
Zo
Zo
Bias
Attenuator
(transmission mode)
Attenuator
(Reflection mode)
Diode ON-attenuated
Diode OFF- transmitted

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