Multi carrier equalization by restoration of redundanc y (merry) for adaptive channel shortening in multi-carrier systems

International Journal of Network Security & Its Applications (IJNSA), Vol.5, No.6, November 2013

Multi-carrier Equalization by Restoration of
RedundancY (MERRY) for Adaptive Channel
Shortening in Multi-carrier Systems
Samir Abd Elghafar1,2, Salaheldin M. Diab1, Bassiouny M. Sallam1, Moawad I. Dessouky1,
El-Sayed M. El-Rabaie1, Fathi E. Abd El-Samie1
1Department of Electronics and Electrical Communications, Faculty of Electronic
Engineering, Menoufia University, Menouf, 32952, Egypt.
2Faculty of Engineering,
Department of Electrical Engineering,Jazan University,Jazan, Saudi Arabia

Abstract
This paper proposes a new blind adaptive channel shortening approach for multi-carrier systems. The
performance of the discrete Fourier transform-DMT (DFT-DMT) system is investigated with the proposed
DST-DMT system over the standard carrier serving area (CSA) loop1. Enhanced bit rates demonstrated
and less complexity also involved by the simulation of the DST-DMT system.

Keywords
DFT, DMT, DST.

1. INTRODUCTION
Multi carrier modulation, particularly Discrete Multi-tone (DMT) modulation, is one of the most
prominent modulation methods for high-speed digital communications. For the channel in the
DMT system separated into self- regulating sub-channels. The full amount of bits transmitted
over the channel would be the total amount of the bits transmitted in each sub-channel. The
binary input data are passed onto a set of parallel, independent sub-channels, each of which is
assigned a fixed number of bits during startup or system initialization. Given the measured SNR
of each sub-channel during startup, the number of bits for each sub-channel is then determined
[1]. Each sub-channel is encoded into a quadrature amplitude modulation (QAM) constellation of
the appropriate size. For example, if four bits are assigned to tone i, then tone i will use a 16QAM constellation for encoding. The modulation function, IFFT, converts the encoded binary
data in QAM constellation format from frequency domain to time domain for transmission .The
DMT demodulation regress the time domain signal back to frequency domain data.
A number of v samples is inserted as a guard period at the start of each DMT symbol of to
decrease inter-symbol interference (ISI) at v ≥ L-1, where L is the length of the channel. Intercarrier interference (ICI) can be reduced when the guard period is a cyclic prefix [2]. The guard
period reduces the channel throughput by a factor of N/ (N+ v), where N is both the symbol
length and FFT length. This factor decreases as v turns into large comparative to N, in order that
the presentation failure care for unreasonable. Hence, v is chosen to be relatively small compared
to N. In the field, however, ADSL and VDSL channel impulse responses can exceed N/16
samples [3].
DOI : 10.5121/ijnsa.2013.5606

65


Time-domain equalizer (TEQ) is used for shortening. Accordingly, the DMT equalizer
construction consists of a flow of a single TEQ to shorten the L, a fast Fourier transform (FFT) to
present multi carrier demodulation, and a single-tap frequency-domain equalizer (FEQ) per tone to
correct frequency distortion in the shortened channel. Different DMT TEQ design methods
optimize FIR coefficients based on training data under different criteria. The method of Minimum
Mean Squared Error (MMSE) design is one of the beneficial methods [4,5,6,7] to minimize the
mean square error among the output of the route consisting of the channel and TEQ filter and the
output of a controlled rout consisting of a transmission delay and a purpose filter. The method of
maximum shortening SNR (MSSNR) [3] deals with a challenge to reduce inter-symbol
interference (ISI) in the time domain. The MSSNR method aims at increasing the quotient of the
energy of the used channel impulse response within an object window of ν +1 samples to that
outer the object window. The Minimum-ISI (Min-ISI) method works by weighting the ISI in the
frequency domain [4], for example, to set the ISI in vacant and noisy sub-channels. The traditional
joint method of TEQ-FEQ structure equalizes all sub-channels that possibly bound bit rate
presentation. An alternate receiver architecture has been proposed in [8], in which to relocate the
TEQ procedures to the FEQ. The joint TEQ-FEQ would provide a multi-tap FEQ where each subchannel is discretely equalized. Another alternative structure has been proposed in [9], in which
the authors proposed the FEQ transfer to the TEQ to give up complex-valued time-domain
equalizer filter banks. Joint equalization methods give up higher data rates for the downstream
ADSL task [10].
Multi-carrier modulation (MCM) techniques including discrete multi-tone (DMT) and orthogonal
frequency division multiplexing (OFDM) have been used extensively in many wireless network
standards, such as IEEE 802.1 la, IEEE 802.16a, and wire-line digital communication systems,
such as (ADSL) [11,12,13]. MCMs based on discrete Fourier transform (DFT). They utilize the
complex exponential functions group such as orthogonal source. Particularly, in discrete multitone (DMT) systems, modulation by the IFFT and demodulation by the FFT create orthogonal
sub-channels [14]. But, the exponential functions are not the rare orthogonal source so as to
construct baseband multicarrier signals. A single trigonometric function can be used to perform as
an orthogonal source to apply to the MCM system, and this system can be synthesized using a
discrete sine transform (DST) [12].
TEQ design methods have more complications, and need a channel approximation. The proposed
algorithm in [15] which is a blind adaptive channel shortening algorithm solves the complexity
problem. This algorithm has low complexity and is globally convergent with only a small loss in
bit rate [15]. So in this paper, we can use the DST; along with its inverse as a replacement for the
inverse fast Fourier transform / fast Fourier transform (IFFT/FFT) stages in DMT systems in
order to enhance the performance of the algorithm in [15] and achieve a higher bit rate with low
complexity.

2. Scheme Design
The scheme design is shown in Figure 1. Every one of the N frequency bins is modulated by a
QAM signal, while frequently a number of bins are left as unacceptable carriers [4]. Modulation
is performed via an IFFT, and demodulation is accomplished via FFT. The TEQ is used to
decrease the channel length, and the follow-on reduced efficient channel is matched by a
frequency-domain equalizer (FEQ).

66


Figure 1 . DFT system model. IFFT: modulation, parallel to serial (P/S), add cyclic prefix (CP), eliminate
cyclic prefix (xCP) and FFT demodulation.

For the conventional DFT-DMT system, the channel input sequence [ x n , fft ] at the output of the
IFFT block can be presented as [16]

x n , fft =

1 N −1
⋅∑ Xb ⋅e
N b =0

j 2πbn
N

,

n = 0 , 1 , K , N −1

(1)

where X b is the encoded bit stream.
When adding the CP, the end v samples are identical to the first v samples in the symbol, i.e.

x( Mk + i ) = x( Mk + i + N ),

i ∈ { , L , v}
1

(2)

The symbol period is M = N + v and k is the symbol index. Figure 2 shows an example of this,
with N = 8, v= 2, and M = N + v = 10, and the symbol pictured is for k = 0. The received data r is
obtained from x by
Lh

r ( Mk + i ) = ∑ h(l ) ⋅ x( Mk + i − l ) + n( Mk + i )

(3)

l =0

67


Figure 2 . Demonstration of the change in the ISI at the end of the received symbol
and at the received CP .

and the equalized data y is obtained from r by
Lw

y ( Mk + i ) = ∑ w ( j ) ⋅ r ( Mk + i − j )

(4)

j =0

The channel has Lh + 1 taps, the TEQ has Lw + 1 taps and the effective channel c = h ∗ w has

Lc + 1 taps, where Lc = Lh + Lw .

3. Description algorithm of MERRY
Consider the example in Figure 2. The broadcasted samples 2 and 10 are the same. But, at the
receiver, the interfering samples earlier than sample 2 are not the whole identical to their
correspondences before sample 10. If h(2 ), h(3) and h(4 ) were zero, then r (2 ) = r (10) . If we
try to force r (2 ) = r (10 ) , we may force h(2 ) = h(3) = h(4 ) = 0, forcing the channel selected the
same like the CP. The location of the window of v non-zero taps can be varied by comparing
r (3) to r (11) , or r (4 ) to r (12 ) , etc. In general, if the channel length Lh + 1 ≤ v , in that case the
previous sample inside the CP must look like the previous sample inside the symbol. Unit cost
function that reproduce:

[

2

]

J ∆ = E y (Mk + v + ∆ ) − y (Mk + v + N + ∆ ) ,

∆ ∈ {0,L , M − 1}

(5)

where ∆ is the sign synchronization parameter, which correspond to the preferred delay of the
channel-TEQ collection. The choice of ∆ affects the cost function.
A randomization gradient descent of Eq.(5) guides to a sightless, adaptive TEQ, because the
broadcasted data not require be identified. The proposed MERRY algorithm performs a
randomization gradient descent of Eq.(5), with a limitation to keep away from the slight solution
w = 0. The MERRY algorithm is summarized as follows:

For symbol k = 0,1,2,L ,
68


~ (k ) = r (Mk + v + ∆ ) − r (Mk + v + N + ∆ )
r
e (k

)=

w

T

(6)

(k )~ (k )
r

(7)

~
w(k + 1) = w(k ) − µe(k )~ ∗ (k )
r

(8)

~
w(k + 1)
w(k + 1) = ~
w(k + 1)

(9)

where r (i ) = [r (i ), r (i − 1),L , r (i − Lw )] , and ∗ denotes complex conjugation.
T

Due to the fact that MERRY compares with CP at the last part of the symbol, one update is
achievable for each symbol. Interchanging applications of the limitation have set up one tap to
unity, retaining a channel approximation and renormalizing to enforce c = 1 instead of w = 1 ,
or with a retribution term in the cost function to put into effect the norm limitation.

4. DST Implementation
Applying the above algorithm to the DST approach solves the complexity problem and achieves
higher bit rates than the DFT system.
For the proposed DST-DMT system, the channel input sequence { x n , DST } at the output of the
IDST block can be presented as [17]
1

x n , DST

 2  2 N −1
 πb
=   ⋅ ∑ Ab ⋅ X b ⋅ sin 
(2n + 1)

 N  b =0
 2N


where
 1

Ab =  2
1


,

for b = 0

n = 0, 1, L , N − 1

(10)

(11)

otherwise

, and X b is the encoded bit stream.
Lh

rDST ( Mk + i ) = ∑ h(l ) ⋅ x DST ( Mk + i − l ) + n( Mk + i )
l =0
Lw

y DST ( Mk + i ) = ∑ w ( j ) ⋅ rDST ( Mk + i − j )
j =0

69


IDST
DST

DST

DST

Figure 3 DST system model.

It is known that the DST has excellent spectral compaction and energy concentration properties
(See Fig.(3)). It uses only real arithmetics as opposed to the complex-valued DFT. This reduces
the signal processing complexity especially for real pulse-amplitude modulation signaling, where
DFT-based processing still uses complex arithmetics and suffers from in-phase/quadrature
imbalance problems which can cause appreciable performance degradation [11-18].
The cost function in Eq. (5) can be rewritten as:

[

2

]

J ∆ , DST = E y DST (Mk + v + ∆ ) − y DST (Mk + v + N + ∆ ) ,

∆ ∈ {0, L , M − 1}

(12)

The MERRY algorithm can be rewritten as:
For symbol k = 0,1,2, L,

~ (k ) = r (Mk + v + ∆ ) − r (Mk + v + N + ∆ )
rDST
DST
DST

(13)

~
e DST (k ) = wT (k )rDST (k )
DST

(14)

~
wDST (k + 1) = wDST (k ) − µe DST (k )~DST (k )
r∗

(15)

~
w (k + 1)
wDST (k + 1) = ~ DST
wDST (k + 1)

(16)

where rDST (i ) = [rDST (i ), rDST (i − 1),L , rDST (i − Lw )] , and ∗ denotes complex conjugation.
T

5. Simulation parameters
We use the standard carrier-serving-area (CSA) loop1 [15] as our test channel. The channel
impulse response consists of 512 samples sampled at a rate of 2.208 MHz. Also employ high pass
filter at cutoff frequency of 5.4 kHz and pass band ripple of 0.5 dB to the standard loop1 to add
the outcome of the splitter by the side of the transmitter. The DC channel (channel 0), channels 15, and the Nyquist channel are not used. The DSL presentation metric is the possible bit rate for a
fixed probability of error,
70


 SNRi 
B = ∑ ln1 +

Γ 

i

(17)

When SNRi = 40 dB in frequency bin i , and Γ is the SNR gap ( Γ =9.8 + 6 - 4.2 = 11.6 dB).
This corresponds to a system margin of 6 dB and a coding gain of 4.2 dB. Both the FFT size and
the DST size are set to N = 512 . We assume that the power allocation is constant over all used
sub-channels and that it is not changed after the TEQ is placed into the system.

6. Simulation Results
A- for DFT Approach
Figure 4 shows simulation results using CSA loop1. The CP size is 32. The FFT length is 512.
We have 16 taps for the TEQ. Figure 5 demonstrates a plan for bit rate opposed versus SNR. The
Bit rate for this plot was computed by running for 5000 symbols and gradually decreasing the step
size over time. For all these SNR values, MERRY approaches the max SSNR solution.

Figure 4. Achievable bit rate vs. time for the DFT system.

71


Figure 5 Achievable bit rate vs. SNR for the DFT system.

B- DST System Results
As compared to figure 4 for the DFT approach, figure 6 shows the achievable bit rate vs.
time for the DST approach at the same simulation parameters. As we can see in the case
of DST approach, the performance is better (high bit rate and low complexity) than that
of the DFT approach.
Also, when comparing figure 5 for the DFT approach with figure 7 for the DST approach,
we can see that the DST approach provides a higher bit rate than the DFT approach at the
same SNR values.

72


Figure 6 Achievable bit rate vs. time for the DST system.

Figure 7 Achievable bit rate vs. SNR for the DST system.

73


8. Conclusions
The proposed IDST/DST has been used for modulation /demodulation stages in DMT systems.
Our simulation results show that the DST-DMT system provides a better performance than the
DFT-DMT system because of the excellent spectral compaction and energy concentration
properties of the DST compared to the DFT. It was concluded that the proposed DST-DMT
system achieves higher bit rates as compared with the conventional DFT-DMT system.

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