Theoretical and experimental analysis of electromagnetic coupling into microwave circuit

International Journal of Electrical and Computer Engineering (IJECE)
Vol. 9, No. 1, February 2019, pp. 181~189
ISSN: 2088-8708, DOI: 10.11591/ijece.v9i1.pp181-189  181
Journal homepage: http://iaescore.com/journals/index.php/IJECE
Theoretical and experimental analysis of electromagnetic
coupling into microwave circuit
Ilham Zerrouk1
, Mohamed Amellal2
, Amine Amharech3
, Mohamed Ramdani4
, Hassane Kabbaj5
1,5Faculty of Sciences and Technologies (FST) Signals, Systems and Components Laboratory (SSCL) Electro-Magnetic
Compatibility (EMC) Unit, Morocco
2,4Ecole Superieure d’Electronique de l’Ouest (ESEO) Electro-Magnetic Compatibility (EMC) Laboratory, France
3National School of Applied Sciences Informatics, Systems, Electrical and Telecommunication Networks Laboratory
(ISETNL) Electronics, Signals and Systems (ESS) Unit, Morocco
Article Info ABSTRACT
Article history:
Received Feb 25, 2018
Revised Jun 30, 2018
Accepted Jul 14, 2018
In this paper, our work is devoted to a time domain analysis of field-to-line
coupling model. The latter is designed with a uniform microstrip
multiconductor transmission line (MTL), connected with a mixed load which
can be linear as a resistance, nonlinear like a diode or complex nonlinear as a
Metal Semiconductor Field-Effect Transistor (MESFET). The finite
difference time-domain technique (FDTD) is used to compute the expression
of voltage and current at the line. The primary advantage of this method over
many existing methods is that nonlinear terminations may be readily
incorporated into the algorithm and the analysis. The numerical predictions
using the proposed method show a good agreement with the GHz Transverse
Electro Magnetic (GTEM) measurement.
Keywords:
EMC
FDTD
Field-to-line coupling
GTEM
Immunity measurements
MESFET
MTL
Copyright © 2019 Institute of Advanced Engineering and Science.
All rights reserved.
Corresponding Author:
Ilham Zerrouk,
Department of Electrical Engineering,
Faculty of Sciences and Technologies,
FSTF BP 2202, Route d'Imouzzer, Fez, Morocco.
Email: ilham.zerrouk@usmba.ac.ma
1. INTRODUCTION
Electro-Magnetic Compatibility (EMC) study has become an important step in the realization of
electronic systems. Indeed, the technological evolutions require more and more integration of the electronic
functions and the electromagnetic coupling effect between external fields and electronic systems.
Furthermore, an increased miniaturization of devices shares the same environment. For these reasons, it is
fundamental to ensure the proper functioning of the various parts contributing in this environment.
However, electromagnetic illumination, attacking the electronic system, can propagate to the
components via the transmission lines, causing the malfunction of the component. In this perspective, the
study of the field-line coupling to external electromagnetic excitation allows an accurate prediction of its
effect. The radiated immunity of electronic systems is usually measured in a GTEM (Transverse
Electromagnetic GHz) cell. It illuminates the tested circuit (CUT) with a linearly polarized plane wave
widely accepted as a sufficient electromagnetic compatibility tool for radiated susceptibility and radiated
emission tests [1].
The analysis of electromagnetic coupling effect has been investigated by various authors and by
several techniques [2]-[7]. In this paper, we investigate the analysis of coupling a uniform lossless MTL
composed of three conductors and an external plane wave. This MTL is linked to nonlinear terminations as

 ISSN: 2088-8708
Int J Elec & Comp Eng, Vol. 9, No. 1, February 2019 : 181- 189
182
the diodes and complex nonlinear loads like the MESFET transistors. The microstrip transmission lines
represent an indispensable element for evaluating the performance of Radio Frequency (RF), high-speed
digital and microwave systems. In fact, RF and microwave integrated circuits nearly always use microstrip
lines for connecting active devices as well as implementing microwave functionality. The MESFET is one of
these devices that offer a superior performance, especially in the region of microwave operations and for use
within RF amplifiers. Besides, MESFETs extend the advantages of field effect transistors to significantly
higher frequencies.
In this paper, we aim to obtain the level of the perturbation generated by an external electromagnetic
field in a microwave circuit. The originality of our work is about the passage of the intrinsic large-signal
model's transistor to a junction diode. This makes easier the insertion of a mixed load at the far end of the
MTL. The method used for the analysis of this structure is the Finite-Difference Time-Domain (FDTD) [8].
It is suitable not only for transient nature problems, but also for the problem of inserting the complex
nonlinear components in the MTL. As well as, the used approach has been proven to be the most powerful
and accurate tool for dealing with problems involving electromagnetic interactions for many reasons [9]-[11].
It is also interesting to mention that this method is less greedy in memory space, more suitable for the
treatment of nonlinear loads and for the resolution of differential equations in the time domain. Hence, the
choice of the FDTD method was driven by its simplicity, flexibility, and robustness.
To illustrate the problem and to direct the rest of the study, we will first discuss in section 2 the
FDTD algorithm for the calculation of voltages and currents along the line, especially at the input and output
with linear termination at both ends of the line. Then, we will give the predictions compared with
measurement results of field-to-line coupling model in section 3, followed by conclusion in section 4.
2. RESEARCH METHOD
3.1. Recursion relations
Consider a lossless uniform planar MTL in quasi-TEM propagation mode. The MTL equations in
the time domain are described by a set of differential equations and the external field is included as a set of
distributed series voltage and parallel current sources along the line. These equations are described by [12]:
𝝏𝑽(𝑧, 𝑡)
𝜕𝑧
+ 𝑳
𝝏𝑰(𝑧, 𝑡)
𝜕𝑡
= 𝑽 𝑭(𝑧, 𝑡)
(1)
𝝏𝑰(𝑧, 𝑡)
𝜕𝑧
+ 𝐂
𝝏𝑽(𝑧, 𝑡)
𝜕𝑡
= 𝑰 𝑭(𝑧, 𝑡)
(2)
with
V (z, t) = E (z, t) −
𝝏𝑬 𝑻(𝑧, 𝑡)
𝜕𝑧
(3)
and
𝑰 (z, t) = −𝐂
𝝏𝑬 𝑻(𝑧, 𝑡)
𝜕𝑡
(4)
Where V(z,t) and I(z,t) are (n × 1) vectors of the total voltages (with respect to the reference conductor) and
currents matrices along the MTL. Furthermore, the VF(z,t) and IF(z,t) are (n×1) vectors of forcing functions
that contain the effects of the incident electromagnetic field. The (n × n) matrices L and C represent the per-
unit-length inductance and capacitance respectively, that characterizes the line.
The terms ET(z, t) and EL(z, t) are (n×1) vectors having the components of the incident electric field
that are transverse to the line and parallel to the line conductors, respectively, with the line conductors
removed [2]. The term ET (z,t) is given by:
𝐸 (𝑧, 𝑡) = 𝐸 ,
(ℎ, 𝑧, 𝑡) − 𝐸 ,
(𝑜, 𝑧, 𝑡) (6)
As already mentioned, in this paper we choose the FDTD method to solve the MTL equations. It is
can be used easily to solve several problems either in the transmission line, or the components of the load. In
addition, it has a high accuracy for a wide variety of radio frequency (RF) and antenna problems.
As it is widely known, in the FDTD approach, both space and time are divided into discrete
segments, the line axis z is discretized in Δz increments and the time variable t is discretized in Δt

Int J Elec & Comp Eng ISSN: 2088-8708 
Theoretical and experimental analysis of electromagnetic coupling into microwave circuit (Ilham Zerrouk)
183
increments. The points of current solutions are evaluated at half-time steps and half-spatial positions [13],
[14] as shown in Figure 1.
Figure 1. Location of the solution variables on the line
The numerical stability of the FDTD technique is assured when [15], [16].
𝒕 ≤
𝑧
𝑣
(7)
where 𝑣 is the maximum velocity of the wave propagation.
After the discretization, the induced voltages and currents can be numerically interpreted at integer-
time steps and integer-spatial positions as:
1
∆𝑧
( 𝑽 − 𝑽 ) = −
𝑳
∆𝑡
𝑰 – 𝑰 −
1
∆𝑧
𝑬 , − 𝑬 , +
1
2
𝑬 ,
+ 𝑬 ,
(8)
1
∆𝑧
𝑰 − 𝑰 = −
𝑪
∆𝑡
( 𝑽 𝒌 − 𝑽 ) + 𝑬 , − 𝑬 ,
(9)
Readjusting (8) and (9) gives the voltage equation at the line as follows:
V = V − E , − E , −


I − I for k=2, 3,…,ndz
(10)
Then, the current equation along the line as:
𝑰 = 𝑰 +

𝑳
𝑬 , − 𝑬 , −

𝑳
𝑽 − 𝑽 + 𝑬 , − 𝑬 , for k=1, 3,…,ndz
(11)
Where we denote
𝑽 ≡ 𝑽[(𝑘 − 1)z, nt] (12a)
𝑰 ≡ 𝑰 𝑘 −
1
2
z, nt
(12b)
𝑬 , ≡ 𝑬 [(𝑘 − 1)z, nt] (13a)
𝑬 , ≡ 𝑬 𝑘 −
1
2
z, nt
(13b)
2.2. Near end voltage
The quantities of interest are the input and output voltages, which can be found by incorporating
boundary conditions. The final updated equation of the line is discretized at the source as:
𝑘 = 1 ; 𝑽 = 𝑽 − 𝑹 𝑰

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We have then:
𝑰 =
𝑽 − 𝑽
𝑹
The currents I1 and Is are separated by
∆𝒛
𝟐
see Figure 1.
Concluding that:
𝑰 =
𝑰 + 𝑰
2
with Rs is the source resistance.
The final expression of the near end voltage is then:
𝑽 = 𝑨 𝟏
𝑩. 𝑽 − 𝑬 , − 𝑬 , +
t
𝑪z
𝑹 (𝑽 + 𝑽 ) − 𝟐𝑰
(14)
2.3. Far end voltage
In this paper, we are interested in the MTL loaded with mixed nonlinear complex loads such as the
MESFETs and diode modeled by a nonlinear capacitance in series with a resistance. We are focuses in the
nonlinear comportment of the MESFET in common-source configuration, whose drain is connected to a
grounded resistor RL. In Figure 2 shown, we adopt his intrinsic large signal taken from [17], [18]:
Figure 2. Equivalent circuit of intrinsic large-signal model of MESFET
Where the nonlinearities in this model are represented by gate-source capacitance and drain-source
current [19], [20] :
𝑪 (𝑽 ) =
𝑪
𝑽 ( )
𝑽
if 𝑉 < 𝑉 (15a)
𝑪 𝑽 =
𝑪
(1 − 𝐹 )
1 −
3
2
𝐹 +
𝑽 (𝑡)
2𝑽
if 𝑽 ≥ (𝑽 . 𝐹 )
(15b)
And
𝑰 = 𝑨 𝟎 𝜹 + 𝑨 𝟏 𝑽 + 𝑨 𝟐 𝑽 + 𝑨 𝟑 𝑽 . tanh(𝜶𝑽 ) (16)
Where Vb is the built-in potential of the Schottky gate and δ is a (n × 1) vector of 1.
Thus a junction diode Dj in series with a 10Ω resistor is used for representing the nonlinear load of
this structure which is modeled by [21]:

185
𝐶 = 𝐶 1 −
𝑉
𝑉
(17)
Where Cj0 is the zero-bias drain-bulk junction capacitance, VJ is the junction built-in potential and MJ is the
grading coefficient. Developing current terms of the MESFET yields:
𝑰 (𝒕) = 𝑪 𝑽 (𝑡)
𝑑𝑽 (𝑡)
𝑑𝑡
+ 𝑪
𝑑𝑽 (𝑡)
𝑑𝑡
−
𝑑𝑽 (𝑡)
𝑑𝑡
(18)
In terms of finite difference, IL can be expressed as:
𝑰 = 𝑪 𝑽
𝑽 − 𝑽
𝑡
+ 𝑪
𝑽 − 𝑽
𝑡
−
𝑽 − 𝑽
𝑡
(19a)
𝑰 = 𝑪 𝑽
𝑽 − 𝑽
𝑡
+ 𝑪
𝑽 − 𝑽
𝑡
−
𝑽 − 𝑽
𝑡
(19b)
The far end voltage is expressed with respect to the current, and is written as follows:
𝑽 = 𝑽 − 𝑬 , − 𝑬 , +
2t
𝑪z
𝑰 −
𝑰 + 𝑰
𝟐
(20)
Equation (21) is obtained by inserting the expressions of 𝑰 and 𝑰 into equation (20):
𝑽 = 𝑽 −
𝑪
𝑪
(𝑽 −𝑽 ) −
𝑪
𝑪
𝑽 − 𝑽 + 𝑽 − 𝑽 +
𝑪
𝑪
(𝑽 − 𝑽 ) +

𝑪
𝑰 − 𝑬 , − 𝑬 ,
(21)
According to the model in Figure 2 we have:
𝑽 = 𝑽 + 𝑹 𝑪 𝑽
𝑽 − 𝑽
t
(22)
Replacing 𝑽 of equation (22) in (21) gives:
𝑽 𝑻 =
𝑪
𝑪
𝑽 +
𝑹 𝑪

𝑽 − 𝑽 − 𝑽 + 𝑬 , − 𝑬 , +
𝑪
𝑪
𝑽 − 𝑽 + 𝑽 − 𝑽 + 𝑽 +
𝑹 𝑪

𝑽 − 𝑽 + 𝑽 − 𝑽 −

𝑪
𝑰
(23)
According to the scheme in Figure 2 we have:
𝑽 𝑻 = −𝑹 𝑳 𝒊 = −𝑹 𝑰 − 𝑰 + 𝑰 + 𝑰 (24)
with
𝑰 =
𝑪 𝒅𝒔
 𝑡
(𝑽 − 𝑽 )
(25)
𝑰 =
𝑪 𝒈𝒅
𝑡
(𝑽 − 𝑽 − 𝑽 + 𝑽 )
(26)
𝑰 = 𝑮 𝒅 𝑽 (27)

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and
𝑰 = 𝑨 𝟎 𝛿 + 𝑨 𝟏 𝑽 + 𝑨 𝟐 ( 𝑽 ) + 𝑨 𝟑 ( 𝑽 ) . tanh(𝜶𝑽 ) (28)
The only unknown term in all these equations is 𝑽 , which can be solved by using the Newton-
Raphson method as shown:
𝑯 ≡ −𝑽 + 𝑓 𝑽 = 0 (29)
Where f is a term of V and where the nonlinear components are collected. The matrix V is found by
solving the following equation:
( 𝑽 ) = ( 𝑽 ) − ( 𝑱 𝑯) 𝑯 (30)
where:
𝑱 𝑯 =
𝑑𝑯
𝑑𝑽
│𝑽 = ( 𝑽 )
Once 𝑽 found, we can determine 𝑽 and 𝑽
3. NUMERICAL AND MEASUREMENT RESULTS
The transient response of the printed circuit board (PCB) multiconductor transmission lines
discussed above will be evaluated by FDTD approach and GTEM measurements. Our case study is a MTL
terminated at the near end of 50Ω resistors (𝑅 = 𝑅 = 𝑅 = 50Ω). Furthermore, this structure is
connected with the arbitrary nonlinear loads as MESFETs, diode and resistances at the far end. In this case
the line is illuminated by an incident electromagnetic field [22], whose incident, azimuth and polarization
angles are respectively: 𝜃 = 90°, 𝜃 = 90°, 𝜑 = −90°.
As an example of results obtained by the presented approach and GTEM cell, we choose three
lossless conductors transmission line see Figure 3. The track width of every conductor is w=3mm are
separated by 1cm and placed on one side of a FR4 substrate having 𝜀 = 4.5, thickness h=1.64mm, and the
total line length is ℓ=3cm.
Knowing these parameters, the per-unit-length inductance and capacitance matrices of the circuit are
computed according to the numerical method giving [23].
0.313116 0.0063895 0.0015995
0.0063895 0.313111 0.0063894
0.0015995 0.0063894 0
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 
 
  
 
 
nH mL
120.276 -0.45951 -0.10285
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/
6
 
 
  
 
 
pF mC
The Figure 3 displays the scheme of our circuit under test. As displayed in the scheme, the nonlinear
complex termination is represented by a MESFET modeled by its intrinsic large-signal model and terminated
of 50Ω. The measurements were performed in the EMC laboratory at the ESEO of Angers-France. As
illumination for our circuit, the GTEM cell was chosen as the plane wave source with an input power
Pin=40dBm and a frequency swept from 50MHz to 4GHz.

187
Figure 3. Circuit under test: Microstrip MTL connected witch MESFETs and diode
The GTEM cell can be treated a characteristically terminated 50Ω waveguide with an input
connector. The Figure 4 shown display the whole equipment, which allows us to carry out the immunity
measurement of our circuit. RF generator for immunity measurements of ESEO-EMC Laboratory as shown
in Figure 5.
Figure 4. Circuit under test in the GTEM
Figure 5. RF generator for immunity measurements of ESEO-EMC Laboratory
As well known the results obtained by the FDTD algorithm are in the time domain, contrariwise to
those obtained with the GTEM cell (frequency), that’s why we used the fast Fourier transform algorithm

 ISSN: 2088-8708
188
(FFT) to compare both results.We mention that our choice of conductors is arbitrary. The Figure 6 shown the
numerical and measurement results of the near ends and output transistor voltages:
In this section, it is explained the results of research and at the same time is given the
comprehensive discussion. Results can be presented in figures, graphs, tables and others that make the reader
understand easily [2], [5]. The discussion can be made in several sub-chapters.
The Figure 6(a) illustrates the voltages induced at the input of the first and third conductors. As we
can notice the voltage reaches its maximum at the value of the rise time of the external wave settled to 1 ns.
Besides, the obtained results are in good agreement with those measured by GTEM cell as shown in
Figure 6(b) The results, displayed in Figure 7, present the effect of the external plane wave to the MESFET
transitor. We observe that the level of the perturbation is nearly 15mV. However, this value can deteriorate
the transistor gain and causing a malfuction not only for the component, but also for the microwave circuit.
(a) (b)
Figure 6. Near end voltage: (a) FDTD approach; (b) GTEM measurements
(a) (b)
Figure 7. Transistor’s output voltage of the 1st
conductor: (a) FDTD approach; (b) GTEM measurements
4. CONCLUSION
This paper presents a technique to conduct a time-domain analysis of field-to-line coupling.
The proposed time-domain code is principally based on discretizing MTL equations by means of FDTD
0.5 1 1.5 2 2.5 3 3.5 4
-0.2
-0.18
-0.16
-0.14
-0.12
-0.1
-0.08
-0.06
-0.04
-0.02
0
Frequency (GHz)
Voltage(V)
V
NE1
[FDTD]
V
NE3
[FDTD]
0.5 1 1.5 2 2.5 3 3.5 4
-0.2
-0.18
-0.16
-0.14
-0.12
-0.1
-0.08
-0.06
-0.04
-0.02
0
Frequency (GHz)
Voltage(V)
V
NE1
[GTEM]
V
NE3
[GTEM]
0 0.5 1 1.5 2 2.5 3 3.5 4
-0.01
-0.005
0
0.005
0.01
0.015
Frequency (GHz)
Voltage(V)
VT1
[FDTD]
0 0.5 1 1.5 2 2.5 3 3.5 4
-0.015
-0.01
-0.005
0
0.005
0.01
0.015
0.02
Frequency (GHz)
Voltage(V)
VT1
[GTEM]

189
approach along with the use of some numerical functions, to determine the electromagnetic coupling effect at
both ends of MTL.
In the order to readily integrate a mixed complex nonlinear load at the end of MTL, we have
implemented a code to switch the intrinsic large signal model’s transistor to a junction diode. The code has
been verified and validated by confronting the obtained results with those carried out with GTEM cell
(IEC 61967–2/18GHz) realized in ESEO-EMC laboratory, which has been widely accepted as a sufficient
tool in electromagnetic compatibility for both radiated susceptibility tests and radiated emission tests.
After several simulations, the developed approach turned out to be simple as well as a powerful
numerical technique. Arbitrary nonlinear complex loads can easily be dealt without major constraints. As a
matter of fact, we treated a lossless multiconductor microstrip line terminated with MESFETs transistor and a
junction diode.
Predictions have been computed for the simulation parameters ∆z=1mm and ∆t=4ps. These results
are compared with those measured by GTEM cell with a frequency swept from 50MHz to 4GHz and an input
power Pin=40dBm. Finally, identical results have been achieved that prove the simplicity and efficiency of
the new time domain technique as well as a good choice of time and space discretization to satisfy the
stability condition in the method.
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