A SAR-Assisted Two-Stage Pipeline ADC.pdf

IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 4, APRIL 2011 859
A SAR-Assisted Two-Stage Pipeline ADC
Chun C. Lee, Member, IEEE, and Michael P. Flynn, Senior Member, IEEE
Abstract—Successive approximation register (SAR) ADC archi-
tectures are popular for achieving high energy efficiency but they
suffer from resolution and speed limitations. On the other hand
pipeline ADC architectures can achieve high resolution and speed
but have lower energy-efficiency and are more complex. We pro-
pose a two-stage pipeline ADC architecture with a large first-stage
resolution, enabled with the help of a SAR-based sub-ADC. The
prototype 12b 50 MS/s ADC achieves an ENOB of 10.4b at Nyquist,
and a figure-of-merit of 52 fJ/conversion-step. The ADC achieves
low-power, high-resolution and high-speed operation without cal-
ibration. The ADC is fabricated in 65 nm and 90 nm CMOS and
occupies a core area of only 0.16 mm2.
Index Terms—Analog-digital conversion, data conversion, low-
power, successive approximation architecture, switched-capacitor
circuits.
I. INTRODUCTION
P ORTABLE electronic applications, including wireless
communication, imaging and video, demand high-reso-
lution ( 12b) and low-distortion analog-to-digital conversion
with a signal bandwidth of tens of MHz. The pipeline ADC
architecture is popular because it achieves both high resolu-
tion and high speed [1]–[3]. However, pipeline ADCs rely
on good component matching and require high-gain and
high-bandwidth op-amps to achieve good performance [4].
Furthermore, aggressive device scaling in modern CMOS tech-
nology, coupled with low supply voltage operation, has made
the design of op-amps difficult. Digital calibration schemes
[5], [6] and comparator-based op-amp techniques [7], [8] have
been proposed to overcome these limitations. Nevertheless,
successive approximation register (SAR) converters have
surpassed pipeline converters in terms of energy efficiency.
Consisting of a comparator, capacitors, switches and control
logic, switched-capacitor SAR ADCs have the advantage of
minimal analog complexity [9]. The simple analog content
makes the SAR architecture well suited to nanometer CMOS
processes [10]–[12], but SAR ADCs are limited in speed due to
their serial decision making process. Furthermore, the effective
resolution of SAR ADCs is limited by comparator noise and
limited capacitor matching [10]–[13]. Moderate resolution
Manuscript received August 24, 2010; revised November 28, 2010; accepted
December 15, 2010. Date of publication February 17, 2011; date of current
version March 25, 2011. This paper was approved by Guest Editor Ajith
Amerasekera.
C. C. Lee is with Intel Corporation, Hillsboro, OR 97124 USA (e-mail:
leechun2000@gmail.com).
M. P. Flynn is with the Department of Electrical and Computer Engineering,
University of Michigan, Ann Arbor, MI 48109-2122 USA.
Color versions of one or more of the figures in this paper are available online
at http://ieeexplore.ieee.org.
Digital Object Identifier 10.1109/JSSC.2011.2108133
SAR ADCs suffer from a large input capacitance because large
unit capacitors are required to achieve good matching [11],
[12].
In this paper, we present the design, analysis, implementa-
tion and prototype measurements of a 12b 50 MS/s SAR-as-
sisted two-stage pipeline ADC [14]. Here, for the first time a
large MDAC resolution ( 4b) is demonstrated in a pipeline
ADC. The prototype ADC consists of a 6b MDAC first stage
and a 7b SAR ADC second stage. A large MDAC resolution,
especially in the first stage of a pipeline, is desired because it
helps to achieve lower power consumption and better linearity
[1], [3]. However, a challenge in implementing a high resolu-
tion MDAC is the high accuracy required for the sub-ADC.
Also, a power-hungry active front-end sample-and-hold (S/H)
is usually necessary when there is a high-resolution first stage
to avoid aperture errors between MDAC sampling and sub-ADC
decision.
The architecture presented here uses the SAR technique
for the sub-ADC of the first-stage 6b MDAC, which both
enables a high stage resolution and eliminates the need for an
active front-end S/H. The first-stage MDAC has a “half-gain”
implementation, which reduces the required op-amp transcon-
ductance required to achieve the same closed-loop op-amp
bandwidth. Thus, it reduces the op-amp power consumption.
The half-gain MDAC also allows lower op-amp gain for
the same differential nonlinearity (DNL). Furthermore, the
half-gain MDAC reduces the op-amp output swing which
makes it possible to achieve larger op-amp gain. All of these
help in eliminating the need for gain calibration. The use of the
SAR architecture in the second stage, enables a large 7b reso-
lution in a single stage, eliminating the need for more pipeline
stages and reducing power. The stage resolutions (6–7b) are
relatively low compared to moderate resolution (9–11b) SAR
ADCs [11], [12], which allows the use of large unit capacitors
with sufficient matching. In this way capacitor mismatch cali-
bration is not needed in the SAR sub-ADCs of both stages. The
prototype 12b 50 MS/s ADC is calibration-free and achieves
64.4 dB signal-to-noise-and-distortion ratio (SNDR) and 75 dB
spurious-free dynamic range (SFDR) near Nyquist. The device
consumes 3.5 mW from a 1.3 V supply, which is equivalent
to a figure-of-merit (FOM) [15] of 52 fJ/conversion-step. The
ADC is implemented in a 65 nm CMOS process and occupies
a core area of 0.16 mm .
Section II analyzes the trade-offs involved in choosing the
first stage resolution of a pipeline ADC. Section III describes
the prototype ADC architecture in detail and discusses its ad-
vantages over conventional pipeline ADC architectures. Finally,
Sections IV and V present circuit details and measured results
of the prototype ADC, respectively.
0018-9200/$26.00 © 2011 IEEE

860 IEEE JOURNAL OF SOLID-STATE CIRCUITS, VOL. 46, NO. 4, APRIL 2011
Fig. 1. Conventional 12b pipeline ADC with 3b stage resolution.
Fig. 2. SC implementation of a 3b MDAC.
II. FIRST STAGE RESOLUTION IN PIPELINE ADCS
A large resolution in the first stage of a pipeline ADC [1], [3]
has been shown to be very beneficial for the performance of the
ADC as a whole. This is because a large resolution in the first
stage reduces power consumption, as well as the noise and non-
linearity contributions of the subsequent stages to the overall
ADC performance. However, a large first stage resolution is
difficult to implement because of the large number of accurate
comparators required to implement a conventional sub-ADC.
Furthermore, an active front-end S/H is needed to reduce aper-
ture and sampling errors when a conventional high resolution
first stage is employed. This section explains the trade-offs in-
volved in choosing an optimum first stage resolution. Fig. 1
shows a block diagram of a conventional 12b pipeline ADC. As
an example, Fig. 2 shows the switched-capacitor circuit imple-
mentation of a 3b (2b effective + 1b redundant) pipeline stage,
also popularly known as the MDAC stage [16].
A. Power Consumption
The power dissipated in a pipeline ADC is dominated by
the front-end stages. This is because the initial stages have the
largest accuracy requirement and thus need to dissipate more
power in their op-amps to achieve accurate settling. The accu-
racy required in the subsequent stages decreases exponentially,
and thus power dissipated in them is low. It has been shown [1],
[3] that a large resolution in the first stage of the pipeline helps
decrease the total power consumption of the ADC. We now dis-
cuss this in detail.

LEE AND FLYNN: A SAR-ASSISTED TWO-STAGE PIPELINE ADC 861
Assuming a first-order step-response, the output of a first
stage MDAC at the end of hold phase (Fig. 2) can be written
as
(1)
where is the time available for settling, is the feedback
factor, is the op-amp transconductance, and is the
total output load of the op-amp.
For every 1-bit increase in first-stage MDAC resolution ,
the feedback factor decreases by a factor of 2.
This also implies a 1-bit decrease in required resolution of
the subsequent stages. Therefore, the poorer feedback factor
is approximately offset by the larger tolerable settling error
. But the output load capacitance also decreases ap-
proximately two-fold.1 This allows for the decrease of op-amp
transconductance which translates to reduction in op-amp
power consumption. The power improvement with the increase
in first-stage MDAC resolution ceases after a point. This hap-
pens because the output load capacitance cannot scale
down further when it becomes dominated by the op-amp’s
output self-parasitics.
B. ADC Linearity
It has been shown [1] that the linearity of a pipeline ADC im-
proves as the first stage resolution increases for two reasons.
First, a high first-stage resolution leads to lower nonlinearity
from capacitor mismatch. Second, the large gain associated with
a high-resolution stage decreases the nonlinearity and noise con-
tributions of the following stages. We now analyze linearity
versus resolution in detail.
DNL due to capacitor mismatch2 in the first-stage of a
pipeline ADC has been shown to be [1]
(2)
where and are the full ADC resolution and first-stage res-
olution in bits, respectively. is the total input sampling
capacitance of the first stage MDAC ( for Fig. 2). We
see that ADC nonlinearity, measured in DNL, due to capacitor
mismatch, improves by a factor of for every 1-bit increase
in first-stage resolution. Furthermore, for every 1-bit increase in
first-stage resolution, the stage-gain increases by a factor of 2.
A larger first-stage gain desensitizes the ADC linearity to the
nonlinearities of the following stages.
On the other hand, the ADC nonlinearity due to the finite gain,
, of the first stage op-amp, does not change with stage resolu-
tion. Pipeline ADC nonlinearity because of first-stage op-amp
gain error, measured as its maximum absolute value of DNL,
1For the same thermal noise and matching, for every 1-bit increase in first
stage resolution, the following stage input capacitance can be decreased by a
factor of 4. The minimum unit capacitor size requirements and presence of
self-parasitics at the op-amp output limit the total load capacitance scaling to
approximately two.
2This is assuming that capacitor mismatch error goes as 1C=C / C ;
and a unary weighted input capacitor array (C in Fig. 2) implementation.
is proportional to the op-amp gain error and number of
quantization steps of the remaining stages . It is thus
given by
(3)
For every 1-bit increase in first-stage resolution , the feed-
back factor decreases by approximately 2. This is
offset by the two-fold increase in the error tolerable at the output
of the MDAC, because the total resolution of the remaining
stage decreases by 1-bit. Thus, nonlinearity, measured
as the ADC’s (in (3)), because of finite op-amp gain
remains the same.
C. Input Sampling Accuracy
During a sampling phase , the input signal is sampled
onto the array of identical capacitors (Fig. 2). During
the previous hold phase , the capacitors store DAC reference
voltage, or 0 V, depending on the sub-ADC decision in
the previous cycle. If the sampling time is small or the set-
tling time constant3 is large, the previous DAC voltages stored
on the capacitors might not be adequately attenuated leading to
sampling errors. We show that the performance degradation of
the sampling circuit caused by these errors decreases as the res-
olution of the first-stage MDAC increases.
To explain this effect, Fig. 3 illustrates input signal sampling
for the first-stage MDAC. For an input sampling capacitor array
of unit capacitors , assuming the input signal is constant
during the sampling period , and assuming simple RC settling,
the potential stored on th capacitor at the end of sampling is
given by
(4)
Here is the input signal we want to settle to, and
is the prior DAC reference voltage stored on the th capacitor.
depends on the sub-ADC output corresponding to the
prior sample and is either or 0 V.
The average of the voltages sampled across all the MDAC
capacitors is given by
(5)
If during the previous hold phase, unit capacitors are con-
nected to and the remaining capacitors are connected
to 0 V, we can write
(6)
Here is the quantized value of the previous input signal
sample and is the associated quantization error.
3 is the product of capacitance C and (source and sampling switch) resis-
tance R.

Fig. 3. Input signal sampling in first-stage MDAC.
TABLE I
SAMPLING CIRCUIT RESOLUTION (IN BITS) FOR DIFFERENT VALUES OF FIRST-STAGE MDAC RESOLUTIONS AND AVAILABLE SETTLING TIMES (T=)
Rewriting (5) we get
(7)
In (7), is a linear function of the previous
sample voltage and causes some low-pass filtering, but no noise
degradation. On the other hand, is a function of
previous sample’s quantization error, which is noise, and thus
causes SNR degradation. Noise degradation can be large for a
small settling time and a small first-stage resolution be-
cause of the error being large.
This noise can be removed by shorting out the sampling ca-
pacitors at the start of the sampling phase [3]. However, this
reduces the effective sampling time which might not be pos-
sible if available sampling time is small. This problem is nearly
absent in pipeline ADCs with a large first-stage resolution be-
cause quantization error is small.
Table I gives the sampling circuit accuracy (in bits) for dif-
ferent values of settling times and first-stage MDAC res-
olutions. We see that for a given , the sampling circuit per-
formance improves by approximately 1b (or 6 dB) for every 1b
increase in the first-stage MDAC resolution.
The sampling inaccuracy discussed in this section pertains to
non flip-around MDAC architectures with 1b redundancy. As
explained in [17], flip-around MDAC architectures with 0.5b re-
dundancy [16] do not suffer from memory-based input sampling
inaccuracy. However, non-flip-around MDAC architecture with
1b redundancy is chosen for this prototype ADC because it has
a lower output swing requirement4 for the op-amp, making it
more suitable for implementation in low voltage scaled CMOS
processes.
D. Sub-ADC Implementation
We now consider the implementation complexity of the
sub-ADC with respect to the stage resolution. The sub-ADC of
4For example, the sub-ADC comparators at 63V =4 thresholds in a 2b
non-flip-around MDAC are removed to form a 1.5b flip around MDAC [16].
This causes the 1.5b MDAC output residue jV j to be V =2 for jV j
3V =4. With no comparator offsets in its sub-ADC, a 2b MDAC output residue
jV j will be V =2.

a pipeline stage quantizes the input signal and sets the bottom
plate switches of the input sampling capacitor array to the ap-
propriate reference voltage for MDAC operation. The sub-ADC
is usually implemented as a low-resolution flash ADC and can
be of low accuracy due to the presence of redundancy [18].
comparators are required to implement an M-bit sub-ADC.
As M increases, the number of comparators required increases
exponentially. Redundancy in the conventional pipeline ADC
architecture allows large comparator offsets provided that the
sub-ADC output is monotonic.5 This translates to a maximum
comparator offset of 1/2LSB, where LSB is resolution of
the sub-ADC. As the stage resolution increases, the LSB size
decreases exponentially, hence the tolerable comparator offset
decreases exponentially. Thus, the number of comparators and
their accuracy required increases exponentially as the pipeline
stage resolution increases, making it difficult to implement
large stage resolutions.
E. Phase Mismatch and Clock Skew Errors
Without an active front-end sample-and-hold, the first-stage
sub-ADC output might not correctly quantize the input signal
stored on the input sampling capacitor array because of phase
mismatch and clock skew [19]. The input sampling capacitor
array and the sub-ADC (Fig. 2) have different input sampling
paths and it is difficult to match the phase delay of these two
paths precisely. Moreover, if the sub-ADC is located far from
the capacitor array, the sampling clock and the sub-ADC deci-
sion clock might be skewed (Fig. 2) causing an aperture error.
This leads to decision errors by the sub-ADC, which can be
large especially for a high-frequency input signal. Redundancy
helps the ADC tolerate decision errors up to 1/2LSB,6 but as
the first-stage resolution increases, the tolerable error decreases
exponentially.
Calibration has been proposed [20] to mitigate these sampling
errors, but most calibration methods calibrate out errors for a
particular input signal frequency and are thus less effective for
other frequencies. Another known method to remove this error
is to use an active S/H at the input of the pipeline ADC. An
active front-end S/H presents a clean sampled-and-held signal
to the input of the first-stage and greatly reduces the effect of
phase mismatch and clock skew. But an active front-end S/H
adds noise and consumes considerable power. Therefore, these
methods are not popular, especially in low-power ADC designs.
III. PROPOSED ADC ARCHITECTURE
The architecture7 we propose in this paper avails of the ad-
vantages of a high-resolution front-end stage and eliminates the
need for several accurate comparators in the sub-ADC and the
need for an active front-end S/H. Fig. 4 shows a block dia-
gram of the proposed ADC architecture. The ADC is a pipeline
5This is assuming that the following stage has a sufficient input signal range
to accommodate and correct such errors.
6This is assuming that the comparators are perfect and have no offsets. Bud-
geting part of the redundancy to comparator offset further reduces the tolerable
phase mismatch and sampling-skew errors.
7A similar SAR sub-ADC architecture was independently developed and pre-
sented in [21] and [22].
Fig. 4. Proposed ADC architecture.
of 6b MDAC and a 7b SAR ADC. The first-stage MDAC has
a 5b effective resolution with 1b redundancy. The first-stage
sub-ADC utilizes the MDAC input sampling capacitor array as
a CDAC to implement a SAR sub-ADC. This SAR sub-ADC
architecture eliminates the disadvantages while retaining the
advantages of a high-resolution front-end pipeline stage. The
first-stage MDAC has a gain of 16 instead of the usual gain of
32 for a 6b MDAC; we name this as a “half-gain” implementa-
tion. The half-gain MDAC reduces the op-amp power consump-
tion and significantly reduces ADC nonlinearity due to finite
op-amp gain. The second stage also employs a SAR architecture
to efficiently implement a large 7b stage resolution eliminating
the need for more pipeline stages. The second-stage also has
a “half-reference” implementation which compensates for the
first-stage half-gain implementation, without requiring an ad-
ditional set of half-reference voltages . This section
analyzes these advantages in detail.
A. SAR Sub-ADC Architecture
Fig. 5 shows the circuit implementation of the 6b first-stage
of the pipeline. The flash sub-ADC of a conventional pipeline
ADC (Fig. 1) is replaced by a SAR sub-ADC. The MDAC input
sampling capacitor array is also utilized by the SAR sub-ADC as
its CDAC. The SAR sub-ADC is integrated into the MDAC by
connecting a single comparator to the top plate of the input ca-
pacitor array. This comparator makes serial decisions and quan-
tizes the signal stored on the capacitor array to 6b, reducing the
required number of comparators from 64 to 1.
The problem of sampling path mismatch, described in
Section II-E, is eliminated because the SAR sub-ADC and
MDAC share the same sampling path, as compared to two
different paths in a flash sub-ADC based MDAC. The single
comparator quantizes the static signal stored on the capacitor
array, hence aperture error is also absent. Thus, this architecture
eliminates the need for an active front-end S/H. The MDAC has
a large 6b resolution, which greatly reduces the noise part of
the sampling error ( in (7)) discussed in Section II-C. This
reduces the performance degradation of the sampling circuit.
A disadvantage of a SAR sub-ADC is the increase in the de-
cision time required for the single comparator to quantize the
input signal to 6b. This increased decision time can reduce the
available sampling time or the hold time of the MDAC. In our
prototype implementation, the MDAC’s sampling time is re-
duced to accommodate the increased decision time. A decrease

Fig. 5. First-stage 6b half-gain MDAC implementation.
in hold time of the MDAC is avoided as it would have neces-
sitated a larger op-amp bandwidth, and larger power consump-
tion. The disadvantage is further mitigated by using a fast com-
parator, which requires only 2.4 ns to make 6-bit decisions. This
comparator is discussed in detail in Section IV-C.
B. Half-Gain MDAC
The operation of the MDAC is at the point where the output
load capacitance (in (1)) is dominated by the op-amp’s
self-parasitics. Thus, the large first-stage 6b resolution helps
achieve the maximum possible reduction in power, as discussed
in Section II-A. To further reduce power consumption we pro-
pose a “half-gain” MDAC architecture. In this implementation
the MDAC gain of 32 (i.e., ) found in conventional archi-
tectures is reduced to 16. This half-gain MDAC helps to fur-
ther reduce the op-amp power consumption and significantly
improves the ADC linearity. We will now discuss these advan-
tages in detail.
The half-gain MDAC implementation helps reduce op-amp
power dissipation because it increases the feedback factor
(in (1)) from 1/33 to 1/17. Thus, the closed-loop bandwidth
is increased approximately two-fold. Equiva-
lently, for the same settling error the
op-amp transconductance can be decreased by a factor of
two.
The half gain implementation also significantly improves the
ADC linearity. This is because of two reasons. First, the almost-
two-fold increase in also halves the ADC’s due
to finite op-amp gain-error (in (3)). Second, since the voltage
swing at the output of the op-amp is decreased by a factor of
two, it allows us to stack more cascode transistors in the op-amp
(Section IV-B) to enhance the op-amp gain (in (3)). This fur-
ther reduces due to first-stage gain-error.
In theory, the decreased voltage swing at the MDAC output
because of the half-gain implementation, is a disadvantage for
the second stage because of the reduced LSB size of the second
stage. However, this is not a significant drawback since 7b SAR
ADCs are usually not thermal-noise-limited and are easy to im-
plement even in this reduced signal range without any signifi-
cant increase in the output load capacitance . Although a
further reduction in the first-stage gain, e.g., a quarter-gain im-
plementation, would be even more advantageous for the first-
stage MDAC, this would lead to a greatly reduced second-stage
LSB size resulting in a thermal-noise-limited design. For this
reason, a larger gain reduction of first-stage MDAC was not im-
plemented to avoid the higher second-stage complexity.
The advantages of the first-stage half-gain MDAC greatly
outweigh any disadvantages to the second-stage implementa-
tion. These advantages do not apply to the use of a half-gain in
low resolution first-stage MDACs (e.g., 1.5b or 2.5b) because
decreasing the 1.5b (2.5b) MDAC gain from 2 (4) to 1 (2) would
lead to a larger noise contribution from the later stages. More-
over, implementing an 11b or 10b ADC is certainly more diffi-
cult than implementing a 7b ADC with the same reduced signal
swing.
C. Second Stage Half-Reference SAR Implementation
The second-stage quantizes the first-stage output to 7b,
within . A relatively long half-clock time-period of
10 ns is available for quantization in this stage. Therefore, SAR
architecture is chosen to implement this large resolution in a
single stage while achieving low power consumption. Fig. 6
shows the implementation of the 7b SAR ADC. As explained
in Section III-B, this ADC needs to quantize its input signal
in a smaller signal range of . In a conventional SAR
ADC implementation, this would require half the reference
voltages of the first-stage . To eliminate the need
for these additional half-reference voltages, we employ the
“half-reference” implementation for the second stage.
In the half-reference implementation, all capacitors are split
into two equal halves. During the sampling phase, the input
signal is sampled onto the entire capacitor array as in a conven-
tional SAR ADC. However, during the decision phase, when
the bottom plates of the capacitors are being switched to the
desired reference voltages, only half of the capacitors are con-
nected to full reference voltage while the other half are
grounded. In this way we get a half-reference implementation
without the actual use of voltages. In this implementa-
tion, the number of required unit capacitors doubles. Thus, the
7b SAR ADC has a CDAC complexity similar to an 8b one. Al-
ternatively, a 7b ADC in half-signal range can be implemented
as an 8b ADC in full-signal range with half of its codes unused.
Therefore, the accuracy requirements of the 7b ADC operating
with half signal range are the same as that for an 8b ADC in full
signal range, which translates to same CDAC requirements in
terms of matching and thermal noise.

Fig. 6. Second-stage 7b SAR ADC with half-reference architecture.
IV. CIRCUIT DETAILS
A. First-Stage 6b Half-Gain MDAC
Fig. 5 shows the circuit details of the first-stage 6b MDAC.
A single-ended version is shown for clarity whereas the actual
implementation is fully differential. Most of the architecture de-
tails of this stage are explained in Sections III-A and III-B. The
array of binary weighted unit capacitors serves as the input
sampling capacitor array for the MDAC as well as the CDAC for
the SAR sub-ADC. The single comparator connected to the top
plate of the capacitor array is the SAR sub-ADC comparator.
This comparator serially quantizes the input signal stored in
the capacitor array. There are large voltage excursions on the
top plate of the capacitor array during the SAR decision phase.
The switch between the top plate of the input capacitor sam-
pling array and the inverting input of the op-amp, controlled
by clock phase , isolates the op-amp from these voltage ex-
cursions. This switch introduces a small parasitic capacitance
at the input of the op-amp that slightly degrades the feedback
factor of the switch-capacitor circuit. This degradation dic-
tates a slight increase in the gain and bandwidth requirements
of the op-amp. The input sampling switches are bootstrapped
for better linearity [23]. The sampling time of 7.6 ns, is reduced
from 10 ns to accommodate SAR decision phase. This sampling
time is sufficient for the sampling circuit to achieve more than
14b precision.
B. First-Stage Op-Amp
Because of 1b redundancy8 and half-gain MDAC implemen-
tation, the first-stage op-amp has a low output single-ended
swing requirement of only 250 mV .9 The low output
swing allows the op-amp to be implemented as an nMOS input
triple-cascode op-amp (Fig. 7). The tail transistor is biased in
the linear region with a V of 70 mV. All other transistors
are biased in the subthreshold region with a constant V of
140 mV. With this biasing scheme, the op-amp supports a
maximum single-ended output swing of 390 mV with a
1.3 V supply. Simulations show that this op-amp has a worst
case gain of 75 dB across process and temperature variations.
A switched capacitor common-mode feedback (CMFB) is used
for this op-amp and the feedback correction applied to the tail
transistor.
C. First-Stage SAR Sub-ADC Comparator
The SAR sub-ADC comparator needs to be fast so as to
have a minimal impact on the sample time available to the
MDAC. Dynamic latches [24], often used in low-power SAR
ADCs [11], [12], need a large decision time ( 200 ps), for
two reasons. First, the stack of four transistors between V
and V ( and , in Fig. 8) slows down
81b redundancy halves the output swing of the stage op-amp, provided the
sub-ADC is perfect and has no offsets.
9To accommodate comparator offsets allowed by redundancy, a larger swing
would be required. For a worst case comparator offset an output swing of
500 mV is required.

Fig. 7. First-stage op-amp.
Fig. 8. Dynamic latch [24].
the regenerative process. Second, the gate voltages of the
input differential pair ( and in Fig. 8) are set by the
input signal and cannot be biased independently of the input
common-mode voltage. This means that we cannot choose a
high input common-mode gate voltage for these nMOS devices
to force a quick decision. Hence, a dynamic latch is not suitable
for this application.
The new comparator introduced here (Fig. 9), is a dynamic
preamplifier-based comparator, which has a small decision time
( 100 ps) and is also low-power. This comparator is a pre-am-
plifier followed by a latch and operates in two phases, and .
In the phase, the preamplifier is switched on and it amplifies
the input voltage onto the input nodes of the latch. The pream-
plifier dissipates 1 mA and has a gain of about 2. The pMOS
loads of the preamplifier are driven into triode region, therefore
the outputs settle very fast with a common-mode voltage near
V . In the phase the pre-amplifier is switched off and the
latch is enabled. The latch has a conventional design with three
transistors stacked between V and V . Careful sizing of the
transistors, coupled with a high initial input voltage, results in
a latch decision time of less than 100 ps. The preamplifier is
switched on and off dynamically, and and are both pulled
low when the comparator is inactive, so that the average power
consumption of the comparator is very low.
The SAR sub-ADC’s CDAC top plate settles during the
phase. The 1b redundancy of the MDAC helps tolerate large
sub-ADC decision errors which relaxes the top plate settling
Fig. 9. Dynamic preamplifier-based comparator for first-stage SAR sub-ADC.
requirements for the SAR sub-ADC. Both the and phases
are each 200 ps long, thus all 6b decisions are achieved in 2.4 ns.
D. First-Stage Clock Generator
An MDAC stage requires non-overlapping clocks to define
its sample and hold phases. Non-overlapping clocks are typi-
cally generated using an SR latch based circuit [25], shown in
Fig. 10(a). The pipeline ADC architecture proposed in this paper
requires additional SAR decision clock phases in between the
sample and hold clock phases. Fig. 10(b) shows the scheme used
to generate all the clock phases required in this new architec-
ture. Delays inserted in the generation of clock help gen-
erate the multiple delayed clock phases, . These clock
phases are then combined to form the high frequency clock
from which SAR comparator clocks ( and in Fig. 9) are
derived. This clocking scheme generates all the necessary clock
phases and also ensures non-overlaps between sample, SAR de-
cision and hold phases.
The end of first-stage sampling, which is the falling edge of
clock in Fig. 5, needs to have low jitter to avoid sampling er-
rors. This proposed clock generator ensures a minimal number
of inverter delays between the falling edge of external clock
CLK and falling edge of . This ensures that little jitter is
added to the input signal sampling. Another advantage of this
clocking scheme is that the SAR decision clock ( and in
Fig. 9) pulsewidth varies with process and temperature changes
tracking the comparator speed.
E. Second-Stage 7b SAR ADC
Fig. 6 shows a single-ended circuit implementation of the
second stage 7b SAR ADC. The actual implementation is fully
differential. This stage has an implementation similar to that of a
conventional SAR ADC. Two unit capacitors are connected in
series to implement a LSB capacitor, reducing the total
number of unit capacitors required from 128 to 67. The 32 1/2
unit capacitors, with their bottom plates grounded during SAR
search phase, implement the half-reference design, as described

Fig. 10. (a) Conventional non-overlapping clock generator. (b) First-stage
clock generator.
in Section III-C. The stage has a sufficient time period of 10 ns to
give a 7b decision. Therefore, a low-power, but slower, dynamic
latch (Fig. 8) is used as its comparator. Despite the half-gain
implementation of first stage, the large front-end gain of 16 al-
most eliminates the entire noise contribution of the second stage.
Thus, capacitor sizing for thermal noise and comparator noise
are of little concern in this stage.
The clocks required for this stage are generated using a cir-
cuit similar to the one used for the first stage [Fig. 10(b)]. Instead
of using the delay generators in a non-overlap clock generator
[Fig. 10(a)], a delay-locked loop (DLL) is used to lock the de-
lays for this stage to a half-clock period of 10 ns. This ensures
a complete 7b decision within the half-clock period available to
this stage.
Fig. 11. 65 nm CMOS prototype die microphotograph.
Fig. 12. Measured INL and DNL at 12b level for 65 nm prototype.
V. MEASURED RESULTS
The prototype ADC is fabricated in one-poly-nine-metal
(1P9M) 65 nm and one-poly-seven-metal (1P7M) 90 nm CMOS
processes. Very similar measurement results were obtained for
the 65 nm and 90 nm prototypes with SNDR differences of less
than 0.5 dB. The only difference in design is that the 65 nm
prototype does not incorporate the half-reference architecture
in its second stage. The core fits within a small footprint of 0.4
mm 0.4 mm, as seen in the 65 nm CMOS die microphotograph
(Fig. 11). “SAMP. CKT.” in Fig. 11 refers to the input sampling
switches and bootstrapping circuit mentioned in Section IV-A.
The ADC accepts a full-scale differential input signal of
2 V and has a differential input capacitive loading of 1 pF
which is small as compared to SAR ADCs of similar or even
lower resolutions [11], [12]. Linearity plots (Fig. 12), measured
at 12b level for the 65 nm prototype clocked at the full con-
version rate of 50 MS/s, show that integral nonlinearity (INL)
is within 1.5LSB and 0.75 LSB. The measured
linearity of 90 nm prototype shows INL of within 1.6LSB
and 0.8 LSB. A peak SNDR of 66 dB (10.7b ENOB)
is measured for a 2 MHz input signal for the 65 nm design.
An 8192 point FFT plot, shown in Fig. 13, demonstrates a
78 dB SFDR for a 4 MHz input signal at 0.5 dB full scale.
Fig. 14 summarizes the measured SFDR and SNDR versus
input frequency. SFDR and SNDR fall to 75 dB and 64.4 dB
respectively for a 20 MHz input signal for both ADCs.

TABLE II
ADC PERFORMANCE SUMMARY
Fig. 13. 8192 point FFT for 4 MHz input (65 nm CMOS design).
Fig. 14. Measured SFDR and SNDR versus input frequency (Fs = 50 MS=s).
The prototype consumes a total power (excluding I/O) of
3.5 mW at the full conversion speed of 50 MS/s. The first stage
op-amp consumes 2.6 mW power and the remaining 0.9 mW is
consumed by the clock generators and by the first and second
stage comparators. The 90 nm ADC consumes slightly larger
total power of 3.6 mW. Table II summarizes the measured spec-
ifications of the prototype ADCs.
VI. CONCLUSION
This paper proposes a new pipeline ADC architecture, which
takes advantage of a large stage resolution in the first stage of the
pipeline. This new architecture incorporates a SAR sub-ADC in
its first-stage, which eliminates the drawbacks associated with
a large resolution pipeline stage. A “half-gain” MDAC is also
proposed, which along with the large resolution, helps reduce
the power consumption of the first-stage op-amp and improves
the ADC linearity. A 7b SAR ADC, implemented as the second-
stage of the ADC, eliminates the need for more pipeline stages
and further reduces the power consumption.
The prototype ADC demonstrates the ability of this new ar-
chitecture. The architecture achieves a calibration-free, high-
resolution, moderate-speed, area-efficient and power-efficient
ADC design, which is difficult to achieve with traditional ADC
architectures such as SAR or pipeline. The prototype design
consumes only 3.5 mW of power, which translates to a low FOM
of 52 fJ/conversion-step.
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Chun C. Lee (S’05–M’10) received the B. Tech.
degree in electronics and communication from IIT
Roorkee, India, in 2002 and the M.S. and Ph.D.
degrees in electrical engineering from the Univer-
sity of Michigan, Ann Arbor, in 2006 and 2009,
respectively.
From 2002 to 2004, he was a Senior Design En-
gineer at Texas Instruments, India, where he helped
design high-speed analog-to-digital converters. He
held a summer internship position at Qualcomm,
San Diego, CA, in 2008, where he was involved
with the design of low-power high-resolution analog-to-digital converters. He
is currently an Analog Engineer with Intel Corporation, Hillsboro, OR. His
interests include mixed-signal and RF circuits.
Dr. Lee received the Institute Silver Medal from IIT Roorkee in 2002.
Michael P. Flynn (S’92–M’95–SM’98) received the
Ph.D. degree from Carnegie Mellon University, Pitts-
burgh, PA, in 1995.
From 1998 to 1991, he was with the National
Microelectronics Research Centre, Cork, Ireland.
He was with National Semiconductor, Santa Clara,
CA, from 1993 to 1995. From 1995 to 1997 he was a
Member of Technical Staff with Texas Instruments,
DSP RD Lab, Dallas, TX. From 1997 to 2001,
he was with Parthus Technologies, Cork. He joined
the University of Michigan, Ann Arbor, in 2001,
where he is currently an Associate Professor. He is Thrust Leader responsible
for Wireless Interfaces at Michigan’s Wireless Integrated Microsystems NSF
Engineering Research Center. His technical interests are in RF circuits, data
conversion, serial transceivers and biomedical systems.
Dr. Flynn is a 2008 Guggenheim Fellow. He received the 2010 College of
Engineering Ted Kennedy Family Team Excellence Award and the 2005–2006
Outstanding Achievement Award from the Department of Electrical Engi-
neering and Computer Science at the University of Michigan. He received
the NSF Early Career Award in 2004. He received the 1992–1993 IEEE
Solid-State Circuits Pre-doctoral Fellowship. He is an Associate Editor of the
IEEE JOURNAL OF SOLID-STATE CIRCUITS (JSSC) and serves on the Technical
Program Committees of the IEEE International Solid-State Circuits Conference
(ISSCC) and the Symposium on VLSI Circuits. He formerly served on the
program committee of the Asian Solid-State Circuits Conference, and was an
Associate Editor of the IEEE TRANSACTIONS ON CIRCUITS AND SYSTEMS II
from 2002 to 2004.

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